Method of signal generation and signal generating device

ABSTRACT

A signal generation method includes phase-changing baseband signals with respective phase changing patterns to generate respective phase-changed signals, each of the phase changing patterns being different from each other, and inverse-fast-Fourier-transforming the phase-changed signals to respective orthogonal frequency division multiplexing (OFDM) transmission signals. Each phase changing pattern has N candidates for an amount of change in a phase, N being an integer greater than two, and each candidate is periodically selected from the N candidates based on subcarriers of the respective OFDM transmission signals, a phase of the respective baseband signals being changed by the each candidate.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.16/239,145, filed Jan. 3, 2019, now U.S. Pat. No. 10,476,270, which is acontinuation of U.S. application Ser. No. 15/987,016, filed May 23,2018, now U.S. Pat. No. 10,225,123, which is a continuation ofapplication Ser. No. 15/496,406, filed Apr. 25, 2017, now U.S. Pat. No.10,009,207, which is a continuation of application Ser. No. 14/501,780,filed Sep. 30, 2014, now U.S. Pat. No. 9,667,333, which is acontinuation of application Ser. No. 13/811,064, now U.S. Pat. No.8,885,596, which is the National Stage of International Application No.PCT/JP2012/000352, filed Jan. 20, 2012, which is based on applicationsNo. 2011-033771 filed Feb. 18, 2011, 2011-051842 filed Mar. 9, 2011,2011-093544 filed Apr. 19, 2011, and 2011-102101 filed Apr. 28, 2011 inJapan. The entire disclosures of the above-identified applications,including the specification, drawings and claims are incorporated hereinby reference in their entirety.

TECHNICAL FIELD

The present invention relates to a transmission device and a receptiondevice for communication using multiple antennas.

BACKGROUND ART

A MIMO (Multiple-Input, Multiple-Output) system is an example of aconventional communication system using multiple antennas. Inmulti-antenna communication, of which the MIMO system is typical,multiple transmission signals are each modulated, and each modulatedsignal is simultaneously transmitted from a different antenna in orderto increase the transmission speed of the data.

FIG. 23 illustrates a sample configuration of a transmission andreception device having two transmit antennas and two receive antennas,and using two transmit modulated signals (transmit streams). In thetransmission device, encoded data are interleaved, the interleaved dataare modulated, and frequency conversion and the like are performed togenerate transmission signals, which are then transmitted from antennas.In this case, the scheme for simultaneously transmitting differentmodulated signals from different transmit antennas at the same time andon a common frequency is a spatial multiplexing MIMO system.

In this context, Patent Literature 1 suggests using a transmissiondevice provided with a different interleaving pattern for each transmitantenna. That is, the transmission device from FIG. 23 should use twodistinct interleaving patterns performed by two interleavers (π_(a) andπ_(b)). As for the reception device, Non-Patent Literature 1 andNon-Patent Literature 2 describe improving reception quality byiteratively using soft values for the detection scheme (by the MIMOdetector of FIG. 23).

As it happens, models of actual propagation environments in wirelesscommunications include NLOS (Non Line-Of-Sight), typified by a Rayleighfading environment is representative, and LOS (Line-Of-Sight), typifiedby a Rician fading environment. When the transmission device transmits asingle modulated signal, and the reception device performs maximal ratiocombination on the signals received by a plurality of antennas and thendemodulates and decodes the resulting signals, excellent receptionquality can be achieved in a LOS environment, in particular in anenvironment where the Rician factor is large. The Rician factorrepresents the received power of direct waves relative to the receivedpower of scattered waves. However, depending on the transmission system(e.g., a spatial multiplexing MIMO system), a problem occurs in that thereception quality deteriorates as the Rician factor increases (seeNon-Patent Literature 3).

FIGS. 24A and 24B illustrate an example of simulation results of the BER(Bit Error Rate) characteristics (vertical axis: BER, horizontal axis:SNR (signal-to-noise ratio) for data encoded with LDPC (low-densityparity-check) codes and transmitted over a 2×2 (two transmit antennas,two receive antennas) spatial multiplexing MIMO system in a Rayleighfading environment and in a Rician fading environment with Ricianfactors of K=3, 10, and 16 dB. FIG. 24A gives the Max-Logapproximation-based log-likelihood ratio (Max-log APP) BERcharacteristics without iterative detection (see Non-Patent Literature 1and Non-Patent Literature 2), while FIG. 24B gives the Max-log APP BERcharacteristic with iterative detection (see Non-Patent Literature 1 andNon-Patent Literature 2) (number of iterations: five). FIGS. 24A and 24Bclearly indicate that, regardless of whether or not iterative detectionis performed, reception quality degrades in the spatial multiplexingMIMO system as the Rician factor increases. Thus, the problem ofreception quality degradation upon stabilization of the propagationenvironment in the spatial multiplexing MIMO system, which does notoccur in a conventional single-modulation signal system, is unique tothe spatial multiplexing MIMO system.

Broadcast or multicast communication is a service applied to variouspropagation environments. The radio wave propagation environment betweenthe broadcaster and the receivers belonging to the users is often a LOSenvironment. When using a spatial multiplexing MIMO system having theabove problem for broadcast or multicast communication, a situation mayoccur in which the received electric field strength is high at thereception device, but in which degradation in reception quality makesservice reception difficult. In other words, in order to use a spatialmultiplexing MIMO system in broadcast or multicast communication in boththe NLOS environment and the LOS environment, a MIMO system that offersa certain degree of reception quality is desirable.

Non-Patent Literature 8 describes a scheme for selecting a codebook usedin precoding (i.e. a precoding matrix, also referred to as a precodingweight matrix) based on feedback information from a communication party.However, Non-Patent Literature 8 does not at all disclose a scheme forprecoding in an environment in which feedback information cannot beacquired from the other party, such as in the above broadcast ormulticast communication.

On the other hand, Non-Patent Literature 4 discloses a scheme forswitching the precoding matrix over time. This scheme is applicable whenno feedback information is available. Non-Patent Literature 4 disclosesusing a unitary matrix as the precoding matrix, and switching theunitary matrix at random, but does not at all disclose a schemeapplicable to degradation of reception quality in the above-describedLOS environment. Non-Patent Literature 4 simply recites hopping betweenprecoding matrices at random. Obviously, Non-Patent Literature 4 makesno mention whatsoever of a precoding method, or a structure of aprecoding matrix, for remedying degradation of reception quality in aLOS environment.

CITATION LIST Patent Literature

[Patent Literature 1]

-   International Patent Application Publication No. WO2005/050885

Non-Patent Literature

[Non-Patent Literature 1]

-   “Achieving near-capacity on a multiple-antenna channel” IEEE    Transaction on communications, vol. 51, no. 3, pp. 389-399, March    2003    [Non-Patent Literature 2]-   “Performance analysis and design optimization of LDPC-coded MIMO    OFDM systems” IEEE Trans. Signal Processing, vol. 52, no. 2, pp.    348-361, February 2004    [Non-Patent Literature 3]-   “BER performance evaluation in 2×2 MIMO spatial multiplexing systems    under Rician fading channels” IEICE Trans. Fundamentals, vol. E91-A,    no. 10, pp. 2798-2807, October 2008    [Non-Patent Literature 4]-   “Turbo space-time codes with time varying linear transformations”    IEEE Trans. Wireless communications, vol. 6, no. 2, pp. 486-493,    February 2007    [Non-Patent Literature 5]-   “Likelihood function for QR-MLD suitable for soft-decision turbo    decoding and its performance” IEICE Trans. Commun., vol. E88-B, no.    1, pp. 47-57, January 2004    [Non-Patent Literature 6]-   “A tutorial on ‘Parallel concatenated (Turbo) coding’, ‘Turbo    (iterative) decoding’ and related topics” IEICE, Technical Report    IT98-51    [Non-Patent Literature 7]-   “Advanced signal processing for PLCs: Wavelet-OFDM” Proc. of IEEE    International symposium on ISPLC 2008, pp. 187-192, 2008    [Non-Patent Literature 8]-   D. J. Love and R. W. Heath Jr., “Limited feedback unitary precoding    for spatial multiplexing systems” IEEE Trans. Inf. Theory, vol. 51,    no. 8, pp. 2967-1976, August 2005    [Non-Patent Literature 9]-   DVB Document A122, Framing structure, channel coding and modulation    for a second generation digital terrestrial television broadcasting    system (DVB-T2), June 2008    [Non-Patent Literature 10]-   L. Vangelista, N. Benvenuto, and S. Tomasin “Key technologies for    next-generation terrestrial digital television standard DVB-T2,”    IEEE Commun. Magazine, vo. 47, no. 10, pp. 146-153, October 2009    [Non-Patent Literature 11]-   T. Ohgane, T. Nishimura, and Y. Ogawa, “Application of space    division multiplexing and those performance in a MIMO channel” IEICE    Trans. Commun., vo. 88-B, no. 5, pp. 1843-1851, May 2005    [Non-Patent Literature 12]-   R. G. Gallager “Low-density parity-check codes,” IRE Trans. Inform.    Theory, IT-8, pp. 21-28, 1962    [Non-Patent Literature 13]-   D. J. C. Mackay, “Good error-correcting codes based on very sparse    matrices,” IEEE Trans. Inform. Theory, vol. 45, no. 2, pp. 399-431,    March 1999.    [Non-Patent Literature 14]-   ETSI EN 302 307, “Second generation framing structure, channel    coding and modulation systems for broadcasting, interactive    services, news gathering and other broadband satellite applications”    v.1.1.2, June 2006    [Non-Patent Literature 15]-   Y.-L. Ueng, and C.-C. Cheng “A fast-convergence decoding method and    memory-efficient VLSI decoder architecture for irregular LDPC codes    in the IEEE 802.16e standards” IEEE VTC-2007 Fall, pp. 1255-1259    [Non-Patent Literature 16]-   S. M. Alamouti “A simple transmit diversity technique for wireless    communications” IEEE J. Select. Areas Commun., vol. 16, no. 8, pp.    1451-1458, October 1998    [Non-Patent Literature 17]-   V. Tarokh, H. Jafrkhani, and A. R. Calderbank “Space-time block    coding for wireless communications: Performance results” IEEE J.    Select. Areas Commun., vol. 17, no. 3, no. 3, pp. 451-460, March    1999

SUMMARY OF INVENTION Technical Problem

An object of the present invention is to provide a MIMO system thatimproves reception quality in a LOS environment.

Solution to Problem

The present invention provides a signal generation method forgenerating, from a plurality of baseband signals, a plurality of signalsfor transmission on a common frequency band and at a common time,comprising the steps of: multiplying a first baseband signal s1generated from a first set of bits by u, and multiplying a secondbaseband signal s2 generated from a second set of bits by v, where u andv denote real numbers different from each other; performing a change ofphase on each of the first baseband signal s1 multiplied by u and thesecond baseband signal s2 multiplied by v, thus generating a firstpost-phase-change baseband signal u×s1′ and a second post-phase-changebaseband signal v×s2′; and applying weighting according to apredetermined matrix F to the first post-phase-change baseband signalu×s1′ and to the second post-phase-change baseband signal v×s2′, thusgenerating the plurality of signals for transmission on the commonfrequency band and at the common time as a first weighted signal z1 anda second weighted signal z2, wherein the first weighted signal z1 andthe second weighted signal z2 satisfy the relation: (z1,z2)^(T)=F(u×s1′, v×s2′)^(T) and the change of phase is performed on thefirst baseband signal s1 multiplied by u and the second baseband signals2 multiplied by v by using a phase modification value sequentiallyselected from among N phase modification value candidates, each of the Nphase modification value candidates being selected at least once withina predetermined period.

The present invention also provides a signal generation apparatus forgenerating, from a plurality of baseband signals, a plurality of signalsfor transmission on a common frequency band and at a common time,comprising: a power changer multiplying a first baseband signal s1generated from a first set of bits by u, and multiplying a secondbaseband signal s2 generated from a second set of bits by v, where u andv denote real numbers different from each other; a phase changerperforming a change of phase on each of the first baseband signal s1multiplied by u and the second baseband signal s2 multiplied by v, thusgenerating a first post-phase-change baseband signal u×s1′ and a secondpost-phase-change baseband signal v×s2′; and a weighting unit applyingweighting according to a predetermined matrix F to the firstpost-phase-change baseband signal u×s1′ and to the secondpost-phase-change baseband signal v×s2′, thus generating the pluralityof signals for transmission on the common frequency band and at thecommon time as a first weighted signal z1 and a second weighted signalz2, wherein the first weighted signal z1 and the second weighted signalz2 satisfy the relation: (z1, z2)^(T)=F(u×s1′, v×s2′)^(T) and the changeof phase is performed on the first baseband signal s1 multiplied by uand the second baseband signal s2 multiplied by v by using a phasemodification value sequentially selected from among N phase modificationvalue candidates, each of the N phase modification value candidatesbeing selected at least once within a predetermined period.

Advantageous Effects of Invention

According to the above structure, the present invention provides asignal generation method and a signal generation apparatus that remedydegradation of reception quality in a LOS environment, thereby providinghigh-quality service to LOS users during broadcast or multicastcommunication.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 illustrates an example of a transmission and reception device ina spatial multiplexing MIMO system.

FIG. 2 illustrates a sample frame configuration.

FIG. 3 illustrates an example of a transmission device applying a phasechanging scheme.

FIG. 4 illustrates another example of a transmission device applying aphase changing scheme.

FIG. 5 illustrates another sample frame configuration.

FIG. 6 illustrates a sample phase changing scheme.

FIG. 7 illustrates a sample configuration of a reception device.

FIG. 8 illustrates a sample configuration of a signal processor in thereception device.

FIG. 9 illustrates another sample configuration of a signal processor inthe reception device.

FIG. 10 illustrates an iterative decoding scheme.

FIG. 11 illustrates sample reception conditions.

FIG. 12 illustrates a further example of a transmission device applyinga phase changing scheme.

FIG. 13 illustrates yet a further example of a transmission deviceapplying a phase changing scheme.

FIGS. 14A and 14B illustrate a further sample frame configuration.

FIGS. 15A and 15B illustrate yet another sample frame configuration.

FIGS. 16A and 16B illustrate still another sample frame configuration.

FIGS. 17A and 17B illustrate still yet another sample frameconfiguration.

FIGS. 18A and 18B illustrate yet a further sample frame configuration.

FIGS. 19A and 19B illustrate examples of a mapping scheme.

FIGS. 20A and 20B illustrate further examples of a mapping scheme.

FIG. 21 illustrates a sample configuration of a weighting unit.

FIG. 22 illustrates a sample symbol rearrangement scheme.

FIG. 23 illustrates another example of a transmission and receptiondevice in a spatial multiplexing MIMO system.

FIGS. 24A and 24B illustrate sample BER characteristics.

FIG. 25 illustrates another sample phase changing scheme.

FIG. 26 illustrates yet another sample phase changing scheme.

FIG. 27 illustrates a further sample phase changing scheme.

FIG. 28 illustrates still a further sample phase changing scheme.

FIG. 29 illustrates still yet a further sample phase changing scheme.

FIG. 30 illustrates a sample symbol arrangement for a modulated signalproviding high received signal quality.

FIG. 31 illustrates a sample frame configuration for a modulated signalproviding high received signal quality.

FIG. 32 illustrates another sample symbol arrangement for a modulatedsignal providing high received signal quality.

FIG. 33 illustrates yet another sample symbol arrangement for amodulated signal providing high received signal quality.

FIG. 34 illustrates variation in numbers of symbols and slots needed percoded block when block codes are used.

FIG. 35 illustrates variation in numbers of symbols and slots needed perpair of coded blocks when block codes are used.

FIG. 36 illustrates an overall configuration of a digital broadcastingsystem.

FIG. 37 is a block diagram illustrating a sample receiver.

FIG. 38 illustrates multiplexed data configuration.

FIG. 39 is a schematic diagram illustrating multiplexing of encoded datainto streams.

FIG. 40 is a detailed diagram illustrating a video stream as containedin a PES packet sequence.

FIG. 41 is a structural diagram of TS packets and source packets in themultiplexed data.

FIG. 42 illustrates PMT data configuration.

FIG. 43 illustrates information as configured in the multiplexed data.

FIG. 44 illustrates the configuration of stream attribute information.

FIG. 45 illustrates the configuration of a video display and audiooutput device.

FIG. 46 illustrates a sample configuration of a communications system.

FIGS. 47A and 47B illustrate a variant sample symbol arrangement for amodulated signal providing high received signal quality.

FIGS. 48A and 48B illustrate another variant sample symbol arrangementfor a modulated signal providing high received signal quality.

FIGS. 49A and 49B illustrate yet another variant sample symbolarrangement for a modulated signal providing high received signalquality.

FIGS. 50A and 50B illustrate a further variant sample symbol arrangementfor a modulated signal providing high received signal quality.

FIG. 51 illustrates a sample configuration of a transmission device.

FIG. 52 illustrates another sample configuration of a transmissiondevice.

FIG. 53 illustrates a further sample configuration of a transmissiondevice.

FIG. 54 illustrates yet a further sample configuration of a transmissiondevice.

FIG. 55 illustrates a baseband signal switcher.

FIG. 56 illustrates yet still a further sample configuration of atransmission device.

FIG. 57 illustrates sample operations of a distributor.

FIG. 58 illustrates further sample operations of a distributor.

FIG. 59 illustrates a sample communications system indicating therelationship between base stations and terminals.

FIG. 60 illustrates an example of transmit signal frequency allocation.

FIG. 61 illustrates another example of transmit signal frequencyallocation.

FIG. 62 illustrates a sample communications system indicating therelationship between a base station, repeaters, and terminals.

FIG. 63 illustrates an example of transmit signal frequency allocationwith respect to the base station.

FIG. 64 illustrates an example of transmit signal frequency allocationwith respect to the repeaters.

FIG. 65 illustrates a sample configuration of a receiver and transmitterin the repeater.

FIG. 66 illustrates a signal data format used for transmission by thebase station.

FIG. 67 illustrates yet still another sample configuration of atransmission device.

FIG. 68 illustrates another baseband signal switcher.

FIG. 69 illustrates a weighting, baseband signal switching, and phasechanging scheme.

FIG. 70 illustrates a sample configuration of a transmission deviceusing an OFDM scheme.

FIGS. 71A and 71B illustrate further sample frame configurations.

FIG. 72 illustrates the numbers of slots and phase changing valuescorresponding to a modulation scheme.

FIG. 73 further illustrates the numbers of slots and phase changingvalues corresponding to a modulation scheme.

FIG. 74 illustrates the overall frame configuration of a signaltransmitted by a broadcaster using DVB-T2.

FIG. 75 illustrates two or more types of signals at the same time.

FIG. 76 illustrates still a further sample configuration of atransmission device.

FIG. 77 illustrates an alternate sample frame configuration.

FIG. 78 illustrates another alternate sample frame configuration.

FIG. 79 illustrates a further alternate sample frame configuration.

FIG. 80 illustrates an example of a signal point layout for 16-QAM inthe IQ plane.

FIG. 81 illustrates an example of a signal point layout for QPSK in theIQ plane.

FIG. 82 schematically shows absolute values of a log-likelihood ratioobtained by the reception device.

FIG. 83 schematically shows absolute values of a log-likelihood ratioobtained by the reception device.

FIG. 84 is an example of a structure of a signal processor pertaining toa weighting unit.

FIG. 85 is an example of a structure of the signal processor pertainingto the weighting unit.

FIG. 86 illustrates an example of a signal point layout for 64-QAM inthe IQ plane.

FIG. 87 shows the modulation scheme, the power changing value and thephase changing value to be set at each time.

FIG. 88 shows the modulation scheme, the power changing value and thephase changing value to be set at each time.

FIG. 89 is an example of a structure of the signal processor pertainingto the weighting unit.

FIG. 90 is an example of a structure of the signal processor pertainingto the weighting unit.

FIG. 91 shows the modulation scheme, the power changing value and thephase changing value to be set at each time.

FIG. 92 shows the modulation scheme, the power changing value and thephase changing value to be set at each time.

FIG. 93 is an example of a structure of the signal processor pertainingto the weighting unit.

FIG. 94 illustrates an example of a signal point layout for 16QAM andQPSK in the IQ plane.

FIG. 95 illustrates an example of a signal point layout for 16QAM andQPSK in the IQ plane.

DESCRIPTION OF EMBODIMENTS

Embodiments of the present invention are described below with referenceto the accompanying drawings.

Embodiment 1

The following describes, in detail, a transmission scheme, atransmission device, a reception scheme, and a reception devicepertaining to the present embodiment.

Before beginning the description proper, an outline of transmissionschemes and decoding schemes in a conventional spatial multiplexing MIMOsystem is provided.

FIG. 1 illustrates the structure of an N_(t)×N_(r) spatial multiplexingMIMO system. An information vector z is encoded and interleaved. Theencoded bit vector u=(u₁, . . . u_(Nt)) is obtained as the interleaveoutput. Here, u_(i)=(u_(i1), . . . u_(iM)) (where M is the number oftransmitted bits per symbol). For a transmit vector s=(s₁, . . .S_(Nt)), a received signal s_(i)=map(u_(i)) is found for transmitantenna #i. Normalizing the transmit energy, this is expressible asE{|s_(i)|²}=E_(s)/N_(t) (where E_(s) is the total energy per channel).The receive vector y=(y₁, . . . y_(Nr))^(T) is expressed in formula 1,below.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 1} \right\rbrack & \; \\\begin{matrix}{y = \left( {y_{1},\ldots\;,y_{N_{r}}} \right)^{T}} \\{= {{H_{NtNr}s} + n}}\end{matrix} & \left( {{formula}\mspace{14mu} 1} \right)\end{matrix}$

Here, H_(NtNr) is the channel matrix, n=(n₁, . . . n_(Nr)) is the noisevector, and the average value of n_(i) is zero for independent andidentically distributed (i.i.d) complex Gaussian noise of variance σ².Based on the relationship between transmitted symbols introduced into areceiver and the received symbols, the probability distribution of thereceived vectors can be expressed as formula 2, below, for amulti-dimensional Gaussian distribution.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 2} \right\rbrack & \; \\{{p\left( y \middle| u \right)} = {\frac{1}{\left( {2\pi\;\sigma^{2}} \right)^{N_{r}}}{\exp\left( {{- \frac{1}{2\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} \right)}}} & \left( {{formula}\mspace{14mu} 2} \right)\end{matrix}$

Here, a receiver performing iterative decoding is considered. Such areceiver is illustrated in FIG. 1 as being made up of an outersoft-in/soft-out decoder and a MIMO detector. The log-likelihood ratiovector (L-value) for FIG. 1 is given by formula 3 through formula 5, asfollows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 3} \right\rbrack & \; \\{{L(u)} = \left( {{L\left( u_{1} \right)},\ldots\;,{L\left( u_{N_{t}} \right)}} \right)^{T}} & \left( {{formula}\mspace{14mu} 3} \right) \\\left\lbrack {{Math}.\mspace{11mu} 4} \right\rbrack & \; \\{{L\left( u_{i} \right)} = \left( {{L\left( u_{i\; 1} \right)},\ldots\;,{L\left( u_{iM} \right)}} \right)} & \left( {{formula}\mspace{14mu} 4} \right) \\\left\lbrack {{Math}.\mspace{11mu} 5} \right\rbrack & \; \\{{L\left( u_{ij} \right)} = {\ln\frac{P\left( {u_{ij} = {+ 1}} \right)}{P\left( {u_{ij} = {- 1}} \right)}}} & \left( {{formula}\mspace{14mu} 5} \right)\end{matrix}$(Iterative Detection Scheme)

The following describes the MIMO signal iterative detection performed bythe N_(t)×N_(r) spatial multiplexing MIMO system.

The log-likelihood ratio of u_(min) is defined by formula 6.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 6} \right\rbrack & \; \\{{L\left( u_{mn} \middle| y \right)} = {\ln\frac{P\left( {u_{mn} = \left. {+ 1} \middle| y \right.} \right)}{P\left( {u_{mn} = \left. {- 1} \middle| y \right.} \right)}}} & \left( {{formula}\mspace{14mu} 6} \right)\end{matrix}$

Through application of Bayes' theorem, formula 6 can be expressed asformula 7.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 7} \right\rbrack & \; \\\begin{matrix}{{L\left( u_{mn} \middle| y \right)} = {\ln\frac{{p\left( {\left. y \middle| u_{mn} \right. = {+ 1}} \right)}{{P\left( {u_{mn} = {+ 1}} \right)}/{p(y)}}}{{p\left( {\left. y \middle| u_{mn} \right. = {- 1}} \right)}{{P\left( {u_{mn} = {- 1}} \right)}/{p(y)}}}}} \\{= {{\ln\frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} + {\ln\frac{p\left( {\left. y \middle| u_{mn} \right. = {+ 1}} \right)}{p\left( {\left. y \middle| u_{mn} \right. = {- 1}} \right)}}}} \\{= {{\ln\frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} +}} \\{\ln\frac{\sum\limits_{U_{{mn},{+ 1}}}{{p\left( y \middle| u \right)}{p\left( u \middle| u_{mn} \right)}}}{\sum\limits_{U_{{mn},{- 1}}}{{p\left( y \middle| u \right)}{p\left( u \middle| u_{mn} \right)}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 7} \right)\end{matrix}$

Note that U_(mn, ±1)={u|u_(mn)=±1}. Through the approximation lnΣa_(j)˜max ln a_(j), formula 7 can be approximated as formula 8. Thesymbol ˜ is herein used to signify approximation.

$\begin{matrix}{\mspace{76mu}\left\lbrack {{Math}.\mspace{11mu} 8} \right\rbrack} & \; \\{{L\left( u_{mn} \middle| y \right)} \approx {{\ln\frac{P\left( {u_{mn} = {+ 1}} \right)}{P\left( {u_{mn} = {- 1}} \right)}} + {\max\limits_{{Umn},{+ 1}}\left\{ {{\ln\;{p\left( y \middle| u \right)}} + {P\left( u \middle| u_{mn} \right)}} \right\}} - {\max\limits_{{Umn},{- 1}}\left\{ {{\ln\;{p\left( y \middle| u \right)}} + {P\left( u \middle| u_{mn} \right)}} \right\}}}} & \left( {{formula}\mspace{14mu} 8} \right)\end{matrix}$

In formula 8, P(u|u_(mn)) and ln P(u|u_(mn)) can be expressed asfollows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 9} \right\rbrack & \; \\{{P\left( {u❘u_{mn}} \right)} = {{\prod\limits_{{({ij})} \neq {({mn})}}\;{P\left( u_{ij} \right)}} = {\prod\limits_{{({ij})} \neq {({mn})}}\frac{\exp\left( \frac{u_{ij}{L\left( u_{ij} \right)}}{2} \right)}{\begin{matrix}{{\exp\left( \frac{L\left( u_{ij} \right)}{2} \right)} +} \\{\exp\left( {- \frac{L\left( u_{ij} \right)}{2}} \right)}\end{matrix}}}}} & \left( {{formula}\mspace{14mu} 9} \right) \\{\;\left\lbrack {{Math}.\mspace{14mu} 10} \right\rbrack} & \; \\{\;{{\ln\;{P\left( {u❘u_{mn}} \right)}} = {\left( {\sum\limits_{ij}\;{\ln\;{P\left( u_{ij} \right)}}} \right) - {\ln\;{P\left( u_{mn} \right)}}}}} & \left( {{formula}\mspace{14mu} 10} \right) \\\left\lbrack {{Math}.\mspace{14mu} 11} \right\rbrack & \; \\\begin{matrix}{{\ln\;{P\left( u_{ij} \right)}} = {{\frac{1}{2}u_{ij}{P\left( u_{ij} \right)}} - {\ln\left( {{\exp\left( \frac{L\left( u_{ij} \right)}{2} \right)} +} \right.}}} \\\left. {\exp\left( {- \frac{L\left( u_{ij} \right)}{2}} \right)} \right) \\{\approx {{\frac{1}{2}u_{ij}{L\left( u_{ij} \right)}} - {\frac{1}{2}{{L\left( u_{ij} \right)}}\mspace{14mu}{for}\mspace{14mu}{{L\left( u_{ij} \right)}}}} > 2} \\{= {{\frac{L\left( u_{ij} \right)}{2}}\left( {{u_{ij}{{sign}\left( {L\left( u_{ij} \right)} \right)}} - 1} \right)}}\end{matrix} & \left( {{formula}\mspace{14mu} 11} \right)\end{matrix}$

Note that the log-probability of the formula given in formula 2 can beexpressed as formula 12.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 12} \right\rbrack & \; \\{{\ln\;{P\left( {y❘u} \right)}} = {{{- \frac{N_{r}}{2}}{\ln\left( {2\;\pi\;\sigma^{2}} \right)}} - {\frac{1}{2\;\sigma^{2}}{{y - {{Hs}(u)}}}^{2}}}} & \left( {{formula}\mspace{14mu} 12} \right)\end{matrix}$

Accordingly, given formula 7 and formula 13, the posterior L-value forthe MAP or APP (a posteriori probability) can be can be expressed asfollows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 13} \right\rbrack & \; \\{{L\left( {u_{mn}❘y} \right)} = {\ln\frac{\sum_{U_{{mn},{+ 1}}}{\exp\begin{Bmatrix}\begin{matrix}{- \frac{1}{2\;\sigma^{2}}} \\{{{y - {{Hs}(u)}}}^{2} +}\end{matrix} \\{\sum\limits_{ij}\;{\ln\;{P\left( u_{ij} \right)}}}\end{Bmatrix}}}{\sum_{U_{{mn},{- 1}}}{\exp\begin{Bmatrix}\begin{matrix}{- \frac{1}{2\;\sigma^{2}}} \\{{{y - {{Hs}(u)}}}^{2} +}\end{matrix} \\{\sum\limits_{ij}\;{\ln\;{P\left( u_{ij} \right)}}}\end{Bmatrix}}}}} & \left( {{formula}\mspace{14mu} 13} \right)\end{matrix}$

This is hereinafter termed iterative APP decoding. Also, given formula 8and formula 12, the posterior L-value for the Max-log APP can be can beexpressed as follows.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 14} \right\rbrack} & \; \\{{L\left( {u_{mn}❘y} \right)} \approx {{\max\limits_{{Umn},{+ 1}}\left\{ {\Psi\left( {u,y,{L(u)}} \right)} \right\}} - {\max\limits_{{Umn},{- 1}}\left\{ {\Psi\left( {u,y,{L(u)}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 14} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 15} \right\rbrack} & \; \\{\mspace{79mu}{{\Psi\left( {u,y,{L(u)}} \right)} = {{{- \frac{1}{2\;\sigma^{2}}}{{y - {{Hs}(u)}}}^{2}} + {\sum\limits_{ij}\;{\ln\;{P\left( u_{ij} \right)}}}}}} & \left( {{formula}\mspace{14mu} 15} \right)\end{matrix}$

This is hereinafter referred to as iterative Max-log APP decoding. Assuch, the external information required by the iterative decoding systemis obtainable by subtracting prior input from formula 13 or from formula14.

(System Model)

FIG. 23 illustrates the basic configuration of a system related to thefollowing explanations. The illustrated system is a 2×2 spatialmultiplexing MIMO system having an outer decoder for each of two streamsA and B. The two outer decoders perform identical LDPC encoding(Although the present example considers a configuration in which theouter encoders use LDPC codes, the outer encoders are not restricted tothe use of LDPC as the error-correcting codes. The example may also berealized using other error-correcting codes, such as turbo codes,convolutional codes, or LDPC convolutional codes. Further, while theouter encoders are presently described as individually configured foreach transmit antenna, no limitation is intended in this regard. Asingle outer encoder may be used for a plurality of transmit antennas,or the number of outer encoders may be greater than the number oftransmit antennas. The system also has interleavers (π_(a), π_(b)) foreach of the streams A and B. Here, the modulation scheme is 2^(h)-QAM(i.e., h bits transmitted per symbol).

The receiver performs iterative detection (iterative APP (or Max-logAPP) decoding) of MIMO signals, as described above. The LDPC codes aredecoded using, for example, sum-product decoding.

FIG. 2 illustrates the frame configuration and describes the symbolorder after interleaving. Here, (i_(a),j_(a)) and (i_(b),j_(b)) can beexpressed as follows.[Math. 16](i _(a) ,j _(a))=π_(a)(Ω_(ia,ja) ^(a))  (formula 16)[Math. 17](i _(b) ,j _(b))=π_(b)(Ω_(ib,jb) ^(a))  (formula 17)

Here, i_(a) and i_(b) represent the symbol order after interleaving,j_(a) and j_(b) represent the bit position in the modulation scheme(where j_(a),j_(b)=1, . . . h), π_(a) and π_(b) represent theinterleavers of streams A and B, and Ω_(ia,ja) ^(a) and Q_(ib,jb) ^(b)represent the data order of streams A and B before interleaving. Notethat FIG. 2 illustrates a situation where i_(a) i_(b).

(Iterative Decoding)

The following describes, in detail, the sum-product decoding used indecoding the LDPC codes and the MIMO signal iterative detectionalgorithm, both used by the receiver.

Sum-Product Decoding

A two-dimensional M×N matrix H={H_(mn)} is used as the check matrix forLDPC codes subject to decoding. For the set[1,N]={1, 2 . . . N}, thepartial sets A(m) and B(n) are defined as follows.[Math. 18]A(m)≡{n:H _(mn)=1}  (formula 18)[Math. 19]B(n)≡{m:H _(mn)=1}  (formula 19)

Here, A(m) signifies the set of column indices equal to 1 for row m ofcheck matrix H, while B(n) signifies the set of row indices equal to 1for row n of check matrix H. The sum-product decoding algorithm is asfollows.

Step A-1 (Initialization): For all pairs (m,n) satisfying H_(mn)=1, setthe prior log ratio β_(mn)=1. Set the loop variable (number ofiterations) l_(sum)=1, and set the maximum number of loops l_(sum,max).

Step A-2 (Processing): For all pairs (m,n) satisfying H_(mn)=1 in theorder m=1, 2, . . . M, update the extrinsic value log ratio α_(mn) usingthe following update formula.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 20} \right\rbrack} & \; \\{\alpha_{mn} = {\left( {\prod\limits_{n^{\prime} \in {{A{(m)}}{\backslash n}}}\;{{sign}\left( {\lambda_{n^{\prime}} + \beta_{{mn}^{\prime}}} \right)}} \right) \times {f\left( {\sum\limits_{n^{\prime} \in {{A{(m)}}{\backslash n}}}\;{f\left( {\lambda_{n^{\prime}} + \beta_{{mn}^{\prime}}} \right)}} \right)}}} & \left( {{formula}\mspace{14mu} 20} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 21} \right\rbrack} & \; \\{\mspace{79mu}{{{sign}(x)} \equiv \left\{ \begin{matrix}1 & {x \geq 0} \\{- 1} & {x < 0}\end{matrix} \right.}} & \left( {{formula}\mspace{14mu} 21} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 22} \right\rbrack} & \; \\{\mspace{85mu}{{f(x)} = {\ln\frac{{\exp(x)} + 1}{{\exp(x)} - 1}}}} & \left( {{formula}\mspace{14mu} 22} \right)\end{matrix}$

where f is the Gallager function. λ_(n) can then be computed as follows.

Step A-3 (Column Operations): For all pairs (m,n) satisfying H_(mn)=1 inthe order n=1, 2, . . . N, update the extrinsic value log ratio β_(mn)using the following update formula.

$\begin{matrix}{\;\left\lbrack {{Math}.\mspace{14mu} 23} \right\rbrack} & \; \\{\beta_{mn} = {\sum\limits_{m^{\prime} \in {{\beta{(n)}}\backslash m}}\;\alpha_{m^{\prime}n}}} & \left( {{formula}\mspace{14mu} 23} \right)\end{matrix}$Step A-4 (Log-likelihood Ratio Calculation): For n∈[1,N], thelog-likelihood ratio L_(n) is computed as follows.

$\begin{matrix}{\;\left\lbrack {{Math}.\mspace{14mu} 24} \right\rbrack} & \; \\{L_{n} = {{\sum\limits_{m^{\prime} \in {{\beta{(n)}}\backslash m}}\;\alpha_{m^{\prime}n}} + \lambda_{n}}} & \left( {{formula}\mspace{14mu} 24} \right)\end{matrix}$Step A-5 (Iteration Count): If l_(sum)<l_(sum,max), then l_(sum) isincremented and the process returns to step A-2. Sum-product decodingends when l_(sum)=l_(sum,max).

The above describes one iteration of sum-product decoding operations.Afterward, MIMO signal iterative detection is performed. The variablesm, n, α_(mn), β_(mn), λ_(n), and L_(n) used in the above explanation ofsum-product decoding operations are expressed as m_(a), n_(a), α^(a)_(mana), β^(a) _(mana), λ_(na), and L_(na) for stream A and as m_(b),n_(b), α^(b) _(mbnb), β^(b) _(mbnb), λ_(nb), and L_(nb) for stream B.

(MIMO Signal Iterative Detection)

The following describes the calculation of μ_(n) for MIMO signaliterative detection.

The following formula is derivable from formula 1.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 25} \right\rbrack & \; \\\begin{matrix}{{y(t)} = \left( {{y_{1}(t)},{y_{2}(t)}} \right)^{T}} \\{= {{{H_{22}(t)}{s(t)}} + {n(t)}}}\end{matrix} & \left( {{formula}\mspace{14mu} 25} \right)\end{matrix}$

Given the frame configuration illustrated in FIG. 2, the followingfunctions are derivable from formula 16 and formula 17.[Math. 26]n _(a)=Ω_(ia,ja) ^(a)  (formula 26)[Math. 27]n _(b)=Ω_(ib,jb) ^(b)  (formula 27)

where n_(a),n_(b)∈[1,N]. For iteration k of MIMO signal iterativedetection, the variables λ_(na), L_(na), λ_(nb), and L_(nb) areexpressed as λ_(k,na), L_(k,na), λ_(κ,nb), and L_(k,nb).

Step B-1 (Initial Detection; k=0)

For initial wave detection, μ_(o,na) and μ_(0,nb) are calculated asfollows. For iterative APP decoding:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 28} \right\rbrack & \; \\{\lambda_{0,n_{X}} = {\ln\frac{\sum_{U_{0,n_{X},{+ 1}}}{\exp\begin{Bmatrix}{- \frac{1}{2\;\sigma^{2}}} \\{\begin{matrix}{{y\left( i_{X} \right)} -} \\{{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}\end{matrix}}^{2}\end{Bmatrix}}}{\sum_{U_{0,n_{X},{- 1}}}{\exp\begin{Bmatrix}{- \frac{1}{2\;\sigma^{2}}} \\{\begin{matrix}{{y\left( i_{X} \right)} -} \\{{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}\end{matrix}}^{2}\end{Bmatrix}}}}} & \left( {{formula}\mspace{14mu} 28} \right)\end{matrix}$For iterative Max-log APP decoding:

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 29} \right\rbrack} & \; \\{\lambda_{0,n_{X}} = {{\max\limits_{U_{0,n_{X},{+ 1}}}\left\{ {\Psi\left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} \right\}} - {\max\limits_{U_{0,n_{X},{- 1}}}\left\{ {\Psi\left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 29} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 30} \right\rbrack} & \; \\{\mspace{79mu}{{\Psi\left( {{u\left( i_{X} \right)},{y\left( i_{X} \right)}} \right)} = {{- \frac{1}{2\;\sigma^{2}}}{{{y\left( i_{X} \right)} - {{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}}}^{2}}}} & \left( {{formula}\mspace{14mu} 30} \right)\end{matrix}$

where X=a,b. Next, the iteration count for the MIMO signal iterativedetection is set to l_(mimo)=0, with the maximum iteration count beingl_(mimo,max).

Step B-2 (Iterative Detection; Iteration k): When the iteration count isk, formula 11, formula 13) through formula 15), formula 16), and formula17) can be expressed as formula 31) through formula 34), below. Notethat (X,Y)=(a,b)(b,a). For iterative APP decoding:

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 31} \right\rbrack} & \; \\{\lambda_{k,n_{X}} = {{L_{{k - 1},\Omega_{{iX},{jX}}^{X}}\left( u_{\Omega_{{iX},{jX}}^{X}} \right)} + {\ln\frac{\begin{matrix}{\sum_{U_{k,n_{X},{+ 1}}}\exp} \\\begin{Bmatrix}{- \frac{1}{2\;\sigma^{2}}} \\\begin{matrix}{{\begin{matrix}{{y\left( i_{X} \right)} -} \\{{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}\end{matrix}}^{2} +} \\{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}\end{matrix}\end{Bmatrix}\end{matrix}}{\begin{matrix}{\sum_{U_{k,n_{X},{- 1}}}\exp} \\\begin{Bmatrix}{- \frac{1}{2\;\sigma^{2}}} \\\begin{matrix}{{\begin{matrix}{{y\left( i_{X} \right)} -} \\{{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}\end{matrix}}^{2} +} \\{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}\end{matrix}\end{Bmatrix}\end{matrix}}}}} & \left( {{formula}\mspace{14mu} 31} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 32} \right\rbrack} & \; \\{{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)} = {{\sum\limits_{\underset{\gamma \neq {j\; X}}{\gamma = 1}}^{h}\;{{\frac{L_{{k - 1},\Omega_{{iX},\gamma}^{X}}\left( u_{\Omega_{{iX},\gamma}^{X}} \right)}{2}}\left( {{u_{\Omega_{{iX},\gamma}^{X}}{{sign}\left( {L_{{k - 1},\Omega_{{iX},\gamma}^{X}}\left( u_{\Omega_{{iX},\gamma}^{X}} \right)} \right)}} - 1} \right)}} + {\sum\limits_{\gamma = 1}^{h}{{\frac{L_{{k - 1},\Omega_{{iX},\gamma}^{Y}}\left( u_{\Omega_{{iX},\gamma}^{Y}} \right)}{2}}\left( {{u_{\Omega_{{iX},\gamma}^{Y}}{{sign}\left( {L_{{k - 1},\Omega_{{iX},\gamma}^{Y}}\left( u_{\Omega_{{iX},\gamma}^{Y}} \right)} \right)}} - 1} \right)}}}} & \left( {{formula}\mspace{14mu} 32} \right)\end{matrix}$For iterative Max-log APP decoding:

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 33} \right\rbrack} & \; \\{\lambda_{k,n_{X}} = {{L_{{k - 1},\Omega_{{iX},{jX}}^{X}}\left( u_{\Omega_{{iX},{jX}}^{X}} \right)} + {\max\limits_{U_{k,n_{X},{+ 1}}}\left\{ {\Psi\left( {{u\left( i_{X} \right)},{{y\left( i_{X} \right)}{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}}} \right)} \right\}} - {\max\limits_{U_{k,n_{X},{- 1}}}\left\{ {\Psi\left( {{u\left( i_{X} \right)},{{y\left( i_{X} \right)}{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}}} \right)} \right\}}}} & \left( {{formula}\mspace{14mu} 33} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 34} \right\rbrack} & \; \\{{\Psi\left( {{u\left( i_{X} \right)},{{y\left( i_{X} \right)}{\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}}} \right)} = {{{- \frac{1}{2\;\sigma^{2}}}{\begin{matrix}{{y\left( i_{X} \right)} -} \\{{H_{22}\left( i_{X} \right)}{s\left( {u\left( i_{X} \right)} \right)}}\end{matrix}}^{2}} + {\rho\left( u_{\Omega_{{iX},{jX}}^{X}} \right)}}} & \left( {{formula}\mspace{14mu} 34} \right)\end{matrix}$Step B-3 (Iteration Count and Codeword Estimation) Ifl_(mimo)<l_(mimo,max), then l_(mimo) is incremented and the processreturns to step B-2. When l_(mimo)=l_(mimo,max), an estimated codewordis found, as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 35} \right\rbrack & \; \\{{\hat{u}}_{n_{X}} = \left\{ \begin{matrix}1 & {L_{l_{mimo},n_{X}} \geq 0} \\{- 1} & {L_{l_{mimo},n_{X}} < 0}\end{matrix} \right.} & \left( {{formula}\mspace{14mu} 35} \right)\end{matrix}$

where X=a,b.

FIG. 3 shows a sample configuration of a transmission device 300pertaining to the present Embodiment. An encoder 302A takes information(data) 301A and a frame configuration signal 313 as input (whichincludes the error-correction scheme, coding rate, block length, andother information used by the encoder 302A in error-correction coding ofthe data, such that the scheme designated by the frame configurationsignal 313 is used. The error-correction scheme may be switched). Inaccordance with the frame configuration signal 313, the encoder 302Aperforms error-correction coding, such as convolutional encoding, LDPCencoding, turbo encoding or similar, and outputs encoded data 303A.

An interleaver 304A takes the encoded data 303A and the frameconfiguration signal 313 as input, performs interleaving, i.e.,rearranges the order thereof, and then outputs interleaved data 305A.(Depending on the frame configuration signal 313, the interleavingscheme may be switched.) A mapper 306A takes the interleaved data 305Aand the frame configuration signal 313 as input and performs modulation,such as QPSK (Quadrature Phase Shift Keying), 16-QAM (16-QuadradatureAmplitude Modulation), or 64-QAM (64-Quadradture Amplitude Modulation)thereon, then outputs a baseband signal 307A. (Depending on the frameconfiguration signal 313, the modulation scheme may be switched.)

FIGS. 19A and 19B illustrate an example of a QPSK modulation mappingscheme for a baseband signal made up of an in-phase component I and aquadrature component Q in the IQ plane. For example, as shown in FIG.19A, when the input data are 00, then the output is I=1.0, Q=1.0.Similarly, when the input data are 01, the output is I=−1.0, Q=1.0, andso on. FIG. 19B illustrates an example of a QPSK modulation mappingscheme in the IQ plane differing from FIG. 19A in that the signal pointsof FIG. 19A have been rotated about the origin to obtain the signalpoints of FIG. 19B. Non-Patent Literature 9 and Non-Patent Literature 10describe such a constellation rotation scheme. Alternatively, the CyclicQ Delay described in Non-Patent Literature 9 and Non-Patent Literature10 may also be adopted. An alternate example, distinct from FIGS. 19Aand 19B, is shown in FIGS. 20A and 20B, which illustrate a signal pointlayout for 16-QAM in the IQ plane. The example of FIG. 20A correspondsto FIG. 19A, while that of FIG. 20B corresponds to FIG. 19B.

An encoder 302B takes information (data) 301B and the frameconfiguration signal 313 as input (which includes the error-correctionscheme, coding rate, block length, and other information used by theencoder 302A in error-correction coding of the data, such that thescheme designated by the frame configuration signal 313 is used. Theerror-correction scheme may be switched). In accordance with the frameconfiguration signal 313, the encoder 302B performs error-correctioncoding, such as convolutional encoding, LDPC encoding, turbo encoding orsimilar, and outputs encoded data 303B.

An interleaver 304B takes the encoded data 303B and the frameconfiguration signal 313 as input, performs interleaving, i.e.,rearranges the order thereof, and outputs interleaved data 305B.(Depending on the frame configuration signal 313, the interleavingscheme may be switched.) A mapper 306B takes the interleaved data 305Band the frame configuration signal 313 as input and performs modulation,such as QPSK, 16-QAM, or 64-QAM thereon, then outputs a baseband signal307B. (Depending on the frame configuration signal 313, the modulationscheme may be switched.) A signal processing scheme informationgenerator 314 takes the frame configuration signal 313 as input andaccordingly outputs signal processing scheme information 315. The signalprocessing scheme information 315 designates the fixed precoding matrixto be used, and includes information on the pattern of phase changesused for changing the phase.

A weighting unit 308A takes baseband signal 307A, baseband signal 307B,and the signal processing scheme information 315 as input and, inaccordance with the signal processing scheme information 315, performsweighting on the baseband signals 307A and 307B, then outputs a weightedsignal 309A. The weighting scheme is described in detail, later.

A wireless unit 310A takes weighted signal 309A as input and performsprocessing such as quadrature modulation, band limitation, frequencyconversion, amplification, and so on, then outputs transmit signal 311A.Transmit signal 311A is then output as radio waves by an antenna 312A.

A weighting unit 308B takes baseband signal 307A, baseband signal 307B,and the signal processing scheme information 315 as input and, inaccordance with the signal processing scheme information 315, performsweighting on the baseband signals 307A and 307B, then outputs weightedsignal 316B.

FIG. 21 illustrates the configuration of the weighting units 308A and308B. The area of FIG. 21 enclosed in the dashed line represents one ofthe weighting units. Baseband signal 307A is multiplied by w11 to obtainw11·s1(t), and multiplied by w21 to obtain w21·s1(t). Similarly,baseband signal 307B is multiplied by w12 to obtain w12·s2(t), andmultiplied by w22 to obtain w22·s2(t). Next, z1(t)=w11·s1(t)+w12·s2(t)and z2(t)=w21·s1(t)+w22·s22(t) are obtained. Here, as explained above,s1(t) and s2(t) are baseband signals modulated according to a modulationscheme such as BPSK (Binary Phase Shift Keying), QPSK, 8-PSK (8-PhaseShift Keying), 16-QAM, 32-QAM (32-Quadrature Amplitude Modulation),64-QAM, 256-QAM 16-APSK (16-Amplitude Phase Shift Keying) and so on.

Both weighting units perform weighting using a fixed precoding matrix.The precoding matrix uses, for example, the scheme of formula 36, andsatisfies the conditions of formula 37 or formula 38, all found below.However, this is only an example. The value of α is not restricted toformula 37 and formula 38, and may take on other values, e.g., α=1.

Here, the precoding matrix is:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 36} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\; 0}} \\{\alpha \times e^{j\; 0}} & e^{j\;\pi}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 36} \right)\end{matrix}$

In formula 36, above, α may be given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 37} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 4}{\sqrt{2} + 2}} & \left( {{formula}\mspace{14mu} 37} \right)\end{matrix}$

Alternatively, in formula 36, above, α may be given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 38} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 3 + \sqrt{5}}{\sqrt{2} + 3 - \sqrt{5}}} & \left( {{formula}\mspace{14mu} 38} \right)\end{matrix}$

The precoding matrix is not restricted to that of formula 36, but mayalso be as indicated by formula 39.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 39} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = \begin{pmatrix}a & b \\c & d\end{pmatrix}} & \left( {{formula}\mspace{14mu} 39} \right)\end{matrix}$

In formula 39, let a=Ae^(jδ11), b=Be^(jδ12), c=Ce^(jδ21,) andd=De^(jδ22). Further, one of a, b, c, and d may be zero. For example,the following configurations are possible: (1) a may be zero while b, c,and d are non-zero, (2) b may be zero while a, c, and d are non-zero,(3) c may be zero while a, b, and d are non-zero, or (4) d may be zerowhile a, b, and c are non-zero.

When any of the modulation scheme, error-correcting codes, and thecoding rate thereof are changed, the precoding matrix may also be set,changed, and fixed for use.

A phase changer 317B takes weighted signal 316B and the signalprocessing scheme information 315 as input, then regularly changes thephase of the signal 316B for output. This regular change is a change ofphase performed according to a predetermined phase changing patternhaving a predetermined period (cycle) (e.g., every n symbols (n being aninteger, n≥1) or at a predetermined interval). The details of the phasechanging pattern are explained below, in Embodiment 4.

Wireless unit 310B takes post-phase-change signal 309B as input andperforms processing such as quadrature modulation, band limitation,frequency conversion, amplification, and so on, then outputs transmitsignal 311B. Transmit signal 311B is then output as radio waves by anantenna 312B.

FIG. 4 illustrates a sample configuration of a transmission device 400that differs from that of FIG. 3. The points of difference of FIG. 4from FIG. 3 are described next.

An encoder 402 takes information (data) 401 and the frame configurationsignal 313 as input, and, in accordance with the frame configurationsignal 313, performs error-correction coding and outputs encoded data402.

A distributor 404 takes the encoded data 403 as input, performsdistribution thereof, and outputs data 405A and data 405B. Although FIG.4 illustrates only one encoder, the number of encoders is not limited assuch. The present invention may also be realized using m encoders (mbeing an integer, m≥1) such that the distributor divides the encodeddata created by each encoder into two groups for distribution.

FIG. 5 illustrates an example of a frame configuration in the timedomain for a transmission device according to the present Embodiment.Symbol 500_1 is for notifying the reception device of the transmissionscheme. For example, symbol 500_1 conveys information such as theerror-correction scheme used for transmitting data symbols, the codingrate thereof, and the modulation scheme used for transmitting datasymbols.

Symbol 501_1 is for estimating channel fluctuations for modulated signalz1(t) (where t is time) transmitted by the transmission device. Symbol502_1 is a data symbol transmitted by modulated signal z1(t) as symbolnumber u (in the time domain). Symbol 503_1 is a data symbol transmittedby modulated signal z1(t) as symbol number u+1.

Symbol 501_2 is for estimating channel fluctuations for modulated signalz2(t) (where t is time) transmitted by the transmission device. Symbol502_2 is a data symbol transmitted by modulated signal z2(t) as symbolnumber u (in the time domain). Symbol 503_2 is a data symbol transmittedby modulated signal z1(t) as symbol number u+1.

Here, the symbols of z1(t) and of z2(t) having the same time (identicaltiming) are transmitted from the transmit antenna using the same(shared/common) frequency.

The following describes the relationships between the modulated signalsz1(t) and z2(t) transmitted by the transmission device and the receivedsignals r1(t) and r2(t) received by the reception device.

In FIG. 5, 504#1 and 504#2 indicate transmit antennas of thetransmission device, while 505#1 and 505#2 indicate receive antennas ofthe reception device. The transmission device transmits modulated signalz1(t) from transmit antenna 504#1 and transmits modulated signal z2(t)from transmit antenna 504#2. Here, the modulated signals z1(t) and z2(t)are assumed to occupy the same (shared/common) frequency (bandwidth).The channel fluctuations in the transmit antennas of the transmissiondevice and the antennas of the reception device are h₁₁(t), h₁₂(t),h₂₁(t), and h₂₂(t), respectively. Assuming that receive antenna 505#1 ofthe reception device receives received signal r1(t) and that receiveantenna 505#2 of the reception device receives received signal r2(t),the following relationship holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 40} \right\rbrack & \; \\{\begin{pmatrix}{r\; 1(t)} \\{r\; 2(t)}\end{pmatrix} = {\begin{pmatrix}{h_{11}(t)} & {h_{12}(t)} \\{h_{21}(t)} & {h_{22}(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 40} \right)\end{matrix}$

FIG. 6 pertains to the weighting scheme (precoding scheme) and the phasechanging scheme of the present Embodiment. A weighting unit 600 is acombined version of the weighting units 308A and 308B from FIG. 3. Asshown, stream s1(t) and stream s2(t) correspond to the baseband signals307A and 307B of FIG. 3. That is, the streams s1(t) and s2(t) arebaseband signals made up of an in-phase component I and a quadraturecomponent Q conforming to mapping by a modulation scheme such as QPSK,16-QAM, and 64-QAM. As indicated by the frame configuration of FIG. 6,stream s1(t) is represented as s1(u) at symbol number u, as s1(u+1) atsymbol number u+1, and so forth. Similarly, stream s2(t) is representedas s2(u) at symbol number u, as s2(u+1) at symbol number u+1, and soforth. The weighting unit 600 takes the baseband signals 307A (s1(t))and 307B (s2(t)) as well as the signal processing scheme information 315from FIG. 3 as input, performs weighting in accordance with the signalprocessing scheme information 315, and outputs the weighted signals 309A(z1(t)) and 316B(z2′(t)) from FIG. 3. The phase changer 317B changes thephase of weighted signal 316B(z2′(t)) and outputs post-phase-changesignal 309B(z2(t)).

Here, given vector W1=(w11,w12) from the first row of the fixedprecoding matrix F, z1(t) is expressible as formula 41, below.[Math. 41]z1(t)=W1×(s1(t),s2(t))^(T)  (formula 41)

Similarly, given vector W2=(w21,w22) from the second row of the fixedprecoding matrix F, and letting the phase changing formula applied bythe phase changer by y(t), then z2(t) is expressible as formula 42,below.[Math. 42]z2(t)=y(t)×W2×(s1(t),s2(t))^(T)  (formula 42)

Here, y(t) is a phase changing formula following a predetermined scheme.For example, given a period (cycle) of four and time u, the phasechanging formula is expressible as formula 43, below.[Math. 43]y(u)=e ^(j0)  (formula 43)

Similarly, the phase changing formula for time u+1 may be, for example,as given by formula 44.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 44} \right\rbrack & \; \\{{y\left( {u + 1} \right)} = e^{j\frac{\pi}{2}}} & \left( {{formula}\mspace{14mu} 44} \right)\end{matrix}$

That is, the phase changing formula for time u+k is expressible asformula 45.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 45} \right\rbrack & \; \\{{y\left( {u + k} \right)} = e^{j\frac{k\;\pi}{2}}} & \left( {{formula}\mspace{14mu} 45} \right)\end{matrix}$

Note that formula 43 through formula 45 are given only as an example ofregular phase changing.

The regular change of phase is not restricted to a period (cycle) offour. Improved reception capabilities (the error-correctioncapabilities, to be exact) may potentially be promoted in the receptiondevice by increasing the period (cycle) number (this does not mean thata greater period (cycle) is better, though avoiding small numbers suchas two is likely ideal).

Furthermore, although formula 43 through formula 45, above, represent aconfiguration in which a change in phase is carried out through rotationby consecutive predetermined phases (in the above formula, every π/2),the change in phase need not be rotation by a constant amount, but mayalso be random. For example, in accordance with the predetermined period(cycle) of y(t), the phase may be changed through sequentialmultiplication as shown in formula 46 and formula 47. The key point ofregular phase changing is that the phase of the modulated signal isregularly changed. The degree of phase change is preferably as even aspossible, such as from −π radians to π radians. However, given that thisdescribes a distribution, random changes are also possible.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 46} \right\rbrack} & \; \\\left. e^{j\; 0}\rightarrow\left. e^{j\frac{\pi}{5}}\rightarrow\left. e^{j\frac{2\pi}{5}}\rightarrow\left. e^{j\frac{3\pi}{5}}\rightarrow\left. e^{j\frac{4\pi}{5}}\rightarrow\left. e^{j\;\pi}\rightarrow\left. e^{j\frac{6\pi}{5}}\rightarrow\left. e^{j\frac{7\pi}{5}}\rightarrow\left. e^{j\frac{8\pi}{5}}\rightarrow e^{j\frac{9\pi}{5}} \right. \right. \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 46} \right) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{14mu} 47} \right\rbrack} & \; \\\left. e^{j\frac{\pi}{2}}\rightarrow\left. e^{j\;\pi}\rightarrow\left. e^{j\frac{3\pi}{2}}\rightarrow\left. e^{j\; 2\;\pi}\rightarrow\left. e^{j\frac{\pi}{4}}\rightarrow\left. e^{j\frac{3}{4}\pi}\rightarrow\left. e^{j\frac{5\pi}{4}}\rightarrow e^{j\frac{7\pi}{4}} \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 47} \right)\end{matrix}$

As such, the weighting unit 600 of FIG. 6 performs precoding usingfixed, predetermined precoding weights, and the phase changer 317Bchanges the phase of the signal input thereto while regularly varyingthe phase changing degree.

When a specialized precoding matrix is used in a LOS environment, thereception quality is likely to improve tremendously. However, dependingon the direct wave conditions, the phase and amplitude components of thedirect wave may greatly differ from the specialized precoding matrix,upon reception. The LOS environment has certain rules. Thus, datareception quality is tremendously improved through a regular changeapplied to a transmit signal that obeys those rules. The presentinvention offers a signal processing scheme for improvements in the LOSenvironment.

FIG. 7 illustrates a sample configuration of a reception device 700pertaining to the present embodiment. Wireless unit 703_X receives, asinput, received signal 702_X received by antenna 701_X, performsprocessing such as frequency conversion, quadrature demodulation, andthe like, and outputs baseband signal 704_X.

Channel fluctuation estimator 705_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₁₁ from formula 40, and outputs channelestimation signal 706_1.

Channel fluctuation estimator 705_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₁₂ from formula 40, and outputs channelestimation signal 706_2.

Wireless unit 703_Y receives, as input, received signal 702_Y receivedby antenna 701_X, performs processing such as frequency conversion,quadrature demodulation, and the like, and outputs baseband signal704_Y.

Channel fluctuation estimator 707_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₂₁ from formula 40, and outputs channelestimation signal 708_1.

Channel fluctuation estimator 707_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₂₂ from formula 40, and outputs channelestimation signal 708_2.

A control information decoder 709 receives baseband signal 704_X andbaseband signal 704_Y as input, detects symbol 500_1 that indicates thetransmission scheme from FIG. 5, and outputs a transmission schemeinformation signal 710 for the transmission device.

A signal processor 711 takes the baseband signals 704_X and 704_Y, thechannel estimation signals 706_1, 706_2, 708_1, and 708_2, and thetransmission scheme information signal 710 as input, performs detectionand decoding, and then outputs received data 712_1 and 712_2.

Next, the operations of the signal processor 711 from FIG. 7 aredescribed in detail. FIG. 8 illustrates a sample configuration of thesignal processor 711 pertaining to the present embodiment. As shown, thesignal processor 711 is primarily made up of an inner MIMO detector,soft-in/soft-out decoders, and a coefficient generator. Non-PatentLiterature 2 and Non-Patent Literature 3 describe a scheme of iterativedecoding using this structure. The MIMO system described in Non-PatentLiterature 2 and Non-Patent Literature 3 is a spatial multiplexing MIMOsystem, while the present Embodiment differs from Non-Patent Literature2 and Non-Patent Literature 3 in describing a MIMO system that regularlychanges the phase over time while using the same precoding matrix.Taking the (channel) matrix H(t) of formula 36, then by letting theprecoding weight matrix from FIG. 6 be F (here, a fixed precoding matrixremaining unchanged for a given received signal) and letting the phasechanging formula used by the phase changer from FIG. 6 be Y(t) (here,Y(t) changes over time t), then the receive vectorR(t)=(r1(t),r2(t))^(T) and the stream vector S(t)=(s1(t),s2(t))^(T) thefollowing function is derived:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 48} \right\rbrack & \; \\{{{R(t)} = {{H(t)} \times {Y(t)} \times F \times {S(t)}}}{where}{{Y(t)} = \begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 48} \right)\end{matrix}$

Here, the reception device may use the decoding schemes of Non-PatentLiterature 2 and 3 on R(t) by computing H(t)×Y(t)×F.

Accordingly, the coefficient generator 819 from FIG. 8 takes atransmission scheme information signal 818 (corresponding to 710 fromFIG. 7) indicated by the transmission device (information for specifyingthe fixed precoding matrix in use and the phase changing pattern usedwhen the phase is changed) and outputs a signal processing schemeinformation signal 820.

The inner MIMO detector 803 takes the signal processing schemeinformation signal as input and performs iterative detection anddecoding using the signal and the relationship thereof to formula 48.The operations thereof are described below.

The processor illustrated in FIG. 8 uses a processing scheme, asillustrated by FIG. 10, to perform iterative decoding (iterativedetection). First, detection of one codeword (or one frame) of modulatedsignal (stream) s1 and of one codeword (or one frame) of modulatedsignal (stream) s2 is performed. As a result, the soft-in/soft-outdecoder obtains the log-likelihood ratio of each bit of the codeword (orframe) of modulated signal (stream) s1 and of the codeword (or frame) ofmodulated signal (stream) s2. Next, the log-likelihood ratio is used toperform a second round of detection and decoding. These operations areperformed multiple times (these operations are hereinafter referred toas iterative decoding (iterative detection)). The following explanationscenter on the creation scheme of the log-likelihood ratio of a symbol ata specific time within one frame.

In FIG. 8, a memory 815 takes baseband signal 801X (corresponding tobaseband signal 704_X from FIG. 7), channel estimation signal group 802X(corresponding to channel estimation signals 706_1 and 706_2 from FIG.7), baseband signal 801Y (corresponding to baseband signal 704_Y fromFIG. 7), and channel estimation signal group 802Y (corresponding tochannel estimation signals 708_1 and 708_2 from FIG. 7) as input,executes (computes) H(t)×Y(t)×F from formula 48 in order to performiterative decoding (iterative detection) and stores the resulting matrixas a transformed channel signal group. The memory 815 then outputs theabove-described signals as needed, specifically as baseband signal 816X,transformed channel estimation signal group 817X, baseband signal 816Y,and transformed channel estimation signal group 817Y.

Subsequent operations are described separately for initial detection andfor iterative decoding (iterative detection).

(Initial Detection)

The inner MIMO detector 803 takes baseband signal 801X, channelestimation signal group 802X, baseband signal 801Y, and channelestimation signal group 802Y as input. Here, the modulation scheme formodulated signal (stream) s1 and modulated signal (stream) s2 is takento be 16-QAM.

The inner MIMO detector 803 first computes H(t)×Y(t)×F from the channelestimation signal groups 802X and 802Y, thus calculating a candidatesignal point corresponding to baseband signal 801X. FIG. 11 representssuch a calculation. In FIG. 11, each black dot is a candidate signalpoint in the IQ plane. Given that the modulation scheme is 16-QAM, 256candidate signal points exist. (However, FIG. 11 is only arepresentation and does not indicate all 256 candidate signal points.)Letting the four bits transmitted in modulated signal s1 be b0, b1, b2,and b3 and the four bits transmitted in modulated signal s2 be b4, b5,b6, and b7, candidate signal points corresponding to (b0, b1, b2, b3,b4, b5, b6, b7) are found in FIG. 11. The Euclidean squared distancebetween each candidate signal point and each received signal point 1101(corresponding to baseband signal 801X) is then computed. The Euclidiansquared distance between each point is divided by the noise variance σ².Accordingly, E_(X)(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. Thatis, E_(X) is the Euclidian squared distance between a candidate signalpoint corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a receivedsignal point, divided by the noise variance. Here, each of the basebandsignals and the modulated signals s1 and s2 is a complex signal.

Similarly, the inner MIMO detector 803 computes H(t)×Y(t)×F from thechannel estimation signal groups 802X and 802Y, calculates candidatesignal points corresponding to baseband signal 801Y, computes theEuclidean squared distance between each of the candidate signal pointsand the received signal points (corresponding to baseband signal 801Y),and divides the Euclidean squared distance by the noise variance σ².Accordingly, E_(Y)(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. Thatis, E_(Y) is the Euclidian squared distance between a candidate signalpoint corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a receivedsignal point, divided by the noise variance.

Next, E_(X)(b0, b1, b2, b3, b4, b5, b6, b7)+E_(Y)(b0, b1, b2, b3, b4,b5, b6, b7)=E(b0, b1, b2, b3, b4, b5, b6, b7) is computed.

The inner MIMO detector 803 outputs E(b0, b1, b2, b3, b4, b5, b6, b7) asa signal 804.

Log-likelihood calculator 805A takes the signal 804 as input, calculatesthe log-likelihood of bits b0, b1, b2, and b3, and outputslog-likelihood signal 806A. Note that this log-likelihood calculationproduces the log-likelihood of a bit being 1 and the log-likelihood of abit being 0. The calculation scheme is as shown in formula 28, formula29, and formula 30, and the details are given by Non-Patent Literature 2and 3.

Similarly, log-likelihood calculator 805A takes the signal 804 as input,calculates the log-likelihood of bits b0, b1, b2, and b3, and outputslog-likelihood signal 806B. A deinterleaver (807A) takes log-likelihoodsignal 806A as input, performs deinterleaving corresponding to that ofthe interleaver (the interleaver (304A) from FIG. 3), and outputsdeinterleaved log-likelihood signal 808A.

Similarly, a deinterleaver (807B) takes log-likelihood signal 806B asinput, performs deinterleaving corresponding to that of the interleaver(the interleaver (304B) from FIG. 3), and outputs deinterleavedlog-likelihood signal 808B.

Log-likelihood ratio calculator 809A takes deinterleaved log-likelihoodsignal 808A as input, calculates the log-likelihood ratio of the bitsencoded by encoder 302A from FIG. 3, and outputs log-likelihood ratiosignal 810A.

Similarly, log-likelihood ratio calculator 809B takes deinterleavedlog-likelihood signal 808B as input, calculates the log-likelihood ratioof the bits encoded by encoder 302B from FIG. 3, and outputslog-likelihood ratio signal 810B.

Soft-in/soft-out decoder 811A takes log-likelihood ratio signal 810A asinput, performs decoding, and outputs decoded log-likelihood ratio 812A.

Similarly, soft-in/soft-out decoder 811B takes log-likelihood ratiosignal 810B as input, performs decoding, and outputs decodedlog-likelihood ratio 812B.

(Iterative Decoding (Iterative Detection), k Iterations)

The interleaver (813A) takes the k−1th decoded log-likelihood ratio 812Adecoded by the soft-in/soft-out decoder as input, performs interleaving,and outputs interleaved log-likelihood ratio 814A. Here, theinterleaving pattern used by the interleaver (813A) is identical to thatof the interleaver (304A) from FIG. 3.

Another interleaver (813B) takes the k−1th decoded log-likelihood ratio812B decoded by the soft-in/soft-out decoder as input, performsinterleaving, and outputs interleaved log-likelihood ratio 814B. Here,the interleaving pattern used by the other interleaver (813B) isidentical to that of another interleaver (304B) from FIG. 3.

The inner MIMO detector 803 takes baseband signal 816X, transformedchannel estimation signal group 817X, baseband signal 816Y, transformedchannel estimation signal group 817Y, interleaved log-likelihood ratio814A, and interleaved log-likelihood ratio 814B as input. Here, basebandsignal 816X, transformed channel estimation signal group 817X, basebandsignal 816Y, and transformed channel estimation signal group 817Y areused instead of baseband signal 801X, channel estimation signal group802X, baseband signal 801Y, and channel estimation signal group 802Ybecause the latter cause delays due to the iterative decoding.

The iterative decoding operations of the inner MIMO detector 803 differfrom the initial detection operations thereof in that the interleavedlog-likelihood ratios 814A and 814B are used in signal processing forthe former. The inner MIMO detector 803 first calculates E(b0, b1, b2,b3, b4, b5, b6, b7) in the same manner as for initial detection. Inaddition, the coefficients corresponding to formula 11 and formula 32are computed from the interleaved log-likelihood ratios 814A and 814B.The value of E(b0, b1, b2, b3, b4, b5, b6, b7) is corrected using thecoefficients so calculated to obtain E′(b0, b1, b2, b3, b4, b5, b6, b7),which is output as the signal 804.

Log-likelihood calculator 805A takes the signal 804 as input, calculatesthe log-likelihood of bits b0, b1, b2, and b3, and outputs thelog-likelihood signal 806A. Note that this log-likelihood calculationproduces the log-likelihood of a bit being 1 and the log-likelihood of abit being 0. The calculation scheme is as shown in formula 31 throughformula 35, and the details are given by Non-Patent Literature 2 and 3.

Similarly, log-likelihood calculator 805B takes the signal 804 as input,calculates the log-likelihood of bits b4, b5, b6, and b7, and outputsthe log-likelihood signal 806A. Operations performed by thedeinterleaver onwards are similar to those performed for initialdetection.

While FIG. 8 illustrates the configuration of the signal processor whenperforming iterative detection, this structure is not absolutelynecessary as good reception improvements are obtainable by iterativedetection alone. As long as the components needed for iterativedetection are present, the configuration need not include theinterleavers 813A and 813B. In such a case, the inner MIMO detector 803does not perform iterative detection.

The key point for the present Embodiment is the calculation ofH(t)×Y(t)×F. As shown in Non-Patent Literature 5 and the like, QRdecomposition may also be used to perform initial detection anditerative detection.

Also, as indicated by Non-Patent Literature 11, MMSE (MinimumMean-Square Error) and ZF (Zero-Forcing) linear operations may beperformed based on H(t)×Y(t)×F when performing initial detection.

FIG. 9 illustrates the configuration of a signal processor, unlike thatof FIG. 8, that serves as the signal processor for modulated signalstransmitted by the transmission device from FIG. 4. The point ofdifference from FIG. 8 is the number of soft-in/soft-out decoders. Asoft-in/soft-out decoder 901 takes the log-likelihood ratio signals 810Aand 810B as input, performs decoding, and outputs a decodedlog-likelihood ratio 902. A distributor 903 takes the decodedlog-likelihood ratio 902 as input for distribution. Otherwise, theoperations are identical to those explained for FIG. 8.

As described above, when a transmission device according to the presentEmbodiment using a MIMO system transmits a plurality of modulatedsignals from a plurality of antennas, changing the phase over time whilemultiplying by the precoding matrix so as to regularly change the phaseresults in improvements to data reception quality for a reception devicein a LOS environment where direct waves are dominant, in contrast to aconventional spatial multiplexing MIMO system.

In the present Embodiment, and particularly in the configuration of thereception device, the number of antennas is limited and explanations aregiven accordingly. However, the Embodiment may also be applied to agreater number of antennas. In other words, the number of antennas inthe reception device does not affect the operations or advantageouseffects of the present Embodiment.

Also, although LDPC codes are described as a particular example, thepresent Embodiment is not limited in this manner. Furthermore, thedecoding scheme is not limited to the sum-product decoding example givenfor the soft-in/soft-out decoder. Other soft-in/soft-out decodingschemes, such as the BCJR algorithm, SOVA, and the Max-Log-Map algorithmmay also be used. Details are provided in Non-Patent Literature 6.

In addition, although the present Embodiment is described using asingle-carrier scheme, no limitation is intended in this regard. Thepresent Embodiment is also applicable to multi-carrier transmission.Accordingly, the present Embodiment may also be realized using, forexample, spread-spectrum communications, OFDM (OrthogonalFrequency-Division Multiplexing), SC-FDMA (Single CarrierFrequency-Division Multiple Access), SC-OFDM (Single Carrier OrthogonalFrequency-Division Multiplexing), wavelet OFDM as described inNon-Patent Literature 7, and so on. Furthermore, in the presentEmbodiment, symbols other than data symbols, such as pilot symbols(preamble, unique word, etc) or symbols transmitting controlinformation, may be arranged within the frame in any manner.

The following describes an example in which OFDM is used as amulti-carrier scheme.

FIG. 12 illustrates the configuration of a transmission device usingOFDM. In FIG. 12, components operating in the manner described for FIG.3 use identical reference numbers.

OFDM-related processor 1201A takes weighted signal 309A as input,performs OFDM-related processing thereon, and outputs transmit signal1202A. Similarly, OFDM-related processor 1201B takes post-phase-changesignal 309B as input, performs OFDM-related processing thereon, andoutputs transmit signal 1202A

FIG. 13 illustrates a sample configuration of the OFDM-relatedprocessors 1201A and 1201B and onward from FIG. 12. Components 1301Athrough 1310A belong between 1201A and 312A from FIG. 12, whilecomponents 1301B through 1310B belong between 1201B and 312B.

Serial-to-parallel converter 1302A performs serial-to-parallelconversion on weighted signal 1301A (corresponding to weighted signal309A from FIG. 12) and outputs parallel signal 1303A.

Reorderer 1304A takes parallel signal 1303A as input, performsreordering thereof, and outputs reordered signal 1305A. Reordering isdescribed in detail later.

IFFT (Inverse Fast Fourier Transform) unit 1306A takes reordered signal1305A as input, applies an IFFT thereto, and outputs post-IFFT signal1307A.

Wireless unit 1308A takes post-IFFT signal 1307A as input, performsprocessing such as frequency conversion and amplification, thereon, andoutputs modulated signal 1309A. Modulated signal 1309A is then output asradio waves by antenna 1310A.

Serial-to-parallel converter 1302B performs serial-to-parallelconversion on weighted signal 1301B (corresponding to post-phase-changesignal 309B from FIG. 12) and outputs parallel signal 1303B.

Reorderer 1304B takes parallel signal 1303B as input, performsreordering thereof, and outputs reordered signal 1305B. Reordering isdescribed in detail later.

IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFTthereto, and outputs post-IFFT signal 1307B.

Wireless unit 1308B takes post-IFFT signal 1307B as input, performsprocessing such as frequency conversion and amplification thereon, andoutputs modulated signal 1309B. Modulated signal 1309B is then output asradio waves by antenna 1310A.

The transmission device from FIG. 3 does not use a multi-carriertransmission scheme. Thus, as shown in FIG. 6, the change of phase isperformed to achieve a period (cycle) of four and the post-phase-changesymbols are arranged with respect to the time domain. As shown in FIG.12, when multi-carrier transmission, such as OFDM, is used, then,naturally, precoded post-phase-change symbols may be arranged withrespect to the time domain as in FIG. 3, and this applies to each(sub-)carrier. However, for multi-carrier transmission, the arrangementmay also be in the frequency domain, or in both the frequency domain andthe time domain. The following describes these arrangements.

FIGS. 14A and 14B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13.The frequency axes are made up of (sub-)carriers 0 through 9. Themodulated signals z1 and z2 share common time (timing) and use a commonfrequency band. FIG. 14A illustrates a reordering scheme for the symbolsof modulated signal z1, while FIG. 14B illustrates a reordering schemefor the symbols of modulated signal z2. With respect to the symbols ofweighted signal 1301A input to serial-to-parallel converter 1302A, theassigned ordering is #0, #1, #2, #3, and so on. Here, given that theexample deals with a period (cycle) of four, #0, #1, #2, and #3 areequivalent to one period (cycle). Similarly, #4n, #4n+1, #4n+2, and#4n+3 (n being a non-zero positive integer) are also equivalent to oneperiod (cycle).

As shown in FIG. 14A, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement. Note that the modulated signals z1 and z2 arecomplex signals.

Similarly, with respect to the symbols of weighted signal 1301B input toserial-to-parallel converter 1302B, the assigned ordering is #0, #1, #2,#3, and so on. Here, given that the example deals with a period (cycle)of four, a different change of phase is applied to each of #0, #1, #2,and #3, which are equivalent to one period (cycle). Similarly, adifferent change of phase is applied to each of #4n, #4n+1, #4n+2, and#4n+3 (n being a non-zero positive integer), which are also equivalentto one period (cycle)

As shown in FIG. 14B, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement.

The symbol group 1402 shown in FIG. 14B corresponds to one period(cycle) of symbols when the phase changing scheme of FIG. 6 is used.Symbol #0 is the symbol obtained by using the phase at time u in FIG. 6,symbol #1 is the symbol obtained by using the phase at time u+1 in FIG.6, symbol #2 is the symbol obtained by using the phase at time u+2 inFIG. 6, and symbol #3 is the symbol obtained by using the phase at timeu+3 in FIG. 6. Accordingly, for any symbol #x, symbol #x is the symbolobtained by using the phase at time u in FIG. 6 when x mod 4 equals 0(i.e., when the remainder of x divided by 4 is 0, mod being the modulooperator), symbol #x is the symbol obtained by using the phase at timeu+1 in FIG. 6 when x mod 4 equals 1, symbol #x is the symbol obtained byusing the phase at time u+2 in FIG. 6 when x mod 4 equals 2, and symbol#x is the symbol obtained by using the phase at time u+3 in FIG. 6 whenx mod 4 equals 3.

In the present Embodiment, modulated signal z1 shown in FIG. 14A has notundergone a change of phase.

As such, when using a multi-carrier transmission scheme such as OFDM,and unlike single carrier transmission, symbols may be arranged withrespect to the frequency domain. Of course, the symbol arrangementscheme is not limited to those illustrated by FIGS. 14A and 14B. Furtherexamples are shown in FIGS. 15A, 15B, 16A, and 16B.

FIGS. 15A and 15B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 15A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 15Billustrates a reordering scheme for the symbols of modulated signal z2.FIGS. 15A and 15B differ from FIGS. 14A and 14B in that differentreordering schemes are applied to the symbols of modulated signal z1 andto the symbols of modulated signal z2. In FIG. 15B, symbols #0 through#5 are arranged at carriers 4 through 9, symbols #6 though #9 arearranged at carriers 0 through 3, and this arrangement is repeated forsymbols #10 through #19. Here, as in FIG. 14B, symbol group 1502 shownin FIG. 15B corresponds to one period (cycle) of symbols when the phasechanging scheme of FIG. 6 is used.

FIGS. 16A and 16B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 16A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 16Billustrates a reordering scheme for the symbols of modulated signal z2.FIGS. 16A and 16B differ from FIGS. 14A and 14B in that, while FIGS. 14Aand 14B showed symbols arranged at sequential carriers, FIGS. 16A and16B do not arrange the symbols at sequential carriers. Obviously, forFIGS. 16A and 16B, different reordering schemes may be applied to thesymbols of modulated signal z1 and to the symbols of modulated signal z2as in FIGS. 15A and 15B.

FIGS. 17A and 17B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from those of FIGS. 14A through 16B. FIG. 17A illustrates areordering scheme for the symbols of modulated signal z1 and FIG. 17Billustrates a reordering scheme for the symbols of modulated signal z2.While FIGS. 14A through 16B show symbols arranged with respect to thefrequency axis, FIGS. 17A and 17B use the frequency and time axestogether in a single arrangement.

While FIG. 6 describes an example where a change of phase is performedin a four slot period (cycle), the following example describes an eightslot period (cycle). In FIGS. 17A and 17B, the symbol group 1702 isequivalent to one period (cycle) of symbols when the phase changingscheme is used (i.e., to eight symbols) such that symbol #0 is thesymbol obtained by using the phase at time u, symbol #1 is the symbolobtained by using the phase at time u+1, symbol #2 is the symbolobtained by using the phase at time u+2, symbol #3 is the symbolobtained by using the phase at time u+3, symbol #4 is the symbolobtained by using the phase at time u+4, symbol #5 is the symbolobtained by using the phase at time u+5, symbol #6 is the symbolobtained by using the phase at time u+6, and symbol #7 is the symbolobtained by using the phase at time u+7. Accordingly, for any symbol #x,symbol #x is the symbol obtained by using the phase at time u when x mod8 equals 0, symbol #x is the symbol obtained by using the phase at timeu+1 when x mod 8 equals 1, symbol #x is the symbol obtained by using thephase at time u+2 when x mod 8 equals 2, symbol #x is the symbolobtained by using the phase at time u+3 when x mod 8 equals 3, symbol #xis the symbol obtained by using the phase at time u+4 when x mod 8equals 4, symbol #x is the symbol obtained by using the phase at timeu+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using thephase at time u+6 when x mod 8 equals 6, and symbol #x is the symbolobtained by using the phase at time u+7 when x mod 8 equals 7. In FIGS.17A and 17B four slots along the time axis and two slots along thefrequency axis are used for a total of 4×2=8 slots, in which one period(cycle) of symbols is arranged. Here, given m×n symbols per period(cycle) (i.e., m×n different phases are available for multiplication),then n slots (carriers) in the frequency domain and m slots in the timedomain should be used to arrange the symbols of each period (cycle),such that m>n. This is because the phase of direct waves fluctuatesslowly in the time domain relative to the frequency domain. Accordingly,the present Embodiment performs a regular change of phase that reducesthe influence of steady direct waves. Thus, the phase changing period(cycle) should preferably reduce direct wave fluctuations. Accordingly,m should be greater than n. Taking the above into consideration, usingthe time and frequency domains together for reordering, as shown inFIGS. 17A and 17B, is preferable to using either of the frequency domainor the time domain alone due to the strong probability of the directwaves becoming regular. As a result, the effects of the presentinvention are more easily obtained. However, reordering in the frequencydomain may lead to diversity gain due the fact that frequency-domainfluctuations are abrupt. As such, using the frequency and time domainstogether for reordering is not always ideal.

FIGS. 18A and 18B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 17A and 14B. FIG. 18A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 18Billustrates a reordering scheme for the symbols of modulated signal z2.Much like FIGS. 17A and 17B, FIGS. 18A and 18B illustrate the use of thetime and frequency domains, together. However, in contrast to FIGS. 17Aand 17B, where the frequency domain is prioritized and the time domainis used for secondary symbol arrangement, FIGS. 18A and 18B prioritizethe time domain and use the frequency domain for secondary symbolarrangement. In FIG. 18B, symbol group 1802 corresponds to one period(cycle) of symbols when the phase changing scheme is used.

In FIGS. 17A, 17B, 18A, and 18B, the reordering scheme applied to thesymbols of modulated signal z1 and the symbols of modulated signal z2may be identical or may differ as in FIGS. 15A and 15B. Both approachesallow good reception quality to be obtained. Also, in FIGS. 17A, 17B,18A, and 18B, the symbols may be arranged non-sequentially as in FIGS.16A and 16B. Both approaches allow good reception quality to beobtained.

FIG. 22 indicates frequency on the horizontal axis and time on thevertical axis thereof, and illustrates an example of a symbol reorderingscheme used by the reorderers 1301A and 1301B from FIG. 13 that differsfrom the above. FIG. 22 illustrates a regular phase changing schemeusing four slots, similar to time u through u+3 from FIG. 6. Thecharacteristic feature of FIG. 22 is that, although the symbols arereordered with respect the frequency domain, when read along the timeaxis, a periodic shift of n (n=1 in the example of FIG. 22) symbols isapparent. The frequency-domain symbol group 2210 in FIG. 22 indicatesfour symbols to which the change of phase is applied at time u throughu+3 from FIG. 6.

Here, symbol #0 is obtained through a change of phase at time u, symbol#1 is obtained through a change of phase at time u+1, symbol #2 isobtained through a change of phase at time u+2, and symbol #3 isobtained through a change of phase at time u+3.

Similarly, for frequency-domain symbol group 2220, symbol #4 is obtainedthrough a change of phase at time u, symbol #5 is obtained through achange of phase at time u+1, symbol #6 is obtained through a change ofphase at time u+2, and symbol #7 is obtained through a change of phaseat time u+3.

The above-described change of phase is applied to the symbol at time $1.However, in order to apply periodic shifting in the time domain, thefollowing phase changes are applied to symbol groups 2201, 2202, 2203,and 2204.

For time-domain symbol group 2201, symbol #0 is obtained through achange of phase at time u, symbol #9 is obtained through a change ofphase at time u+1, symbol #18 is obtained through a change of phase attime u+2, and symbol #27 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2202, symbol #28 is obtained through achange of phase at time u, symbol #1 is obtained through a change ofphase at time u+1, symbol #10 is obtained through a change of phase attime u+2, and symbol #19 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2203, symbol #20 is obtained through achange of phase at time u, symbol #29 is obtained through a change ofphase at time u+1, symbol #2 is obtained through a change of phase attime u+2, and symbol #11 is obtained through a change of phase at timeu+3.

For time-domain symbol group 2204, symbol #12 is obtained through achange of phase at time u, symbol #21 is obtained through a change ofphase at time u+1, symbol #30 is obtained through a change of phase attime u+2, and symbol #3 is obtained through a change of phase at timeu+3.

The characteristic feature of FIG. 22 is seen in that, taking symbol #11as an example, the two neighbouring symbols thereof having the same timein the frequency domain (#10 and #12) are both symbols changed using adifferent phase than symbol #11, and the two neighbouring symbolsthereof having the same carrier in the time domain (#2 and #20) are bothsymbols changed using a different phase than symbol #11. This holds notonly for symbol #11, but also for any symbol having two neighboringsymbols in the frequency domain and the time domain. Accordingly, phasechanging is effectively carried out. This is highly likely to improvedate reception quality as influence from regularizing direct waves isless prone to reception.

Although FIG. 22 illustrates an example in which n=1, the invention isnot limited in this manner. The same may be applied to a case in whichn=3. Furthermore, although FIG. 22 illustrates the realization of theabove-described effects by arranging the symbols in the frequency domainand advancing in the time domain so as to achieve the characteristiceffect of imparting a periodic shift to the symbol arrangement order,the symbols may also be randomly (or regularly) arranged to the sameeffect.

Embodiment 2

In Embodiment 1, described above, phase changing is applied to aweighted (precoded with a fixed precoding matrix) signal z(t). Thefollowing Embodiments describe various phase changing schemes by whichthe effects of Embodiment 1 may be obtained.

In the above-described Embodiment, as shown in FIGS. 3 and 6, phasechanger 317B is configured to perform a change of phase on only one ofthe signals output by the weighting unit 600.

However, phase changing may also be applied before precoding isperformed by the weighting unit 600. In addition to the componentsillustrated in FIG. 6, the transmission device may also feature theweighting unit 600 before the phase changer 317B, as shown in FIG. 25.

In such circumstances, the following configuration is possible. Thephase changer 317B performs a regular change of phase with respect tobaseband signal s2(t), on which mapping has been performed according toa selected modulation scheme, and outputs s2′(t)=s2(t)y(t) (where y(t)varies over time t). The weighting unit 600 executes precoding on s2′t,outputs z2(t)=W2 s 2′(t) (see formula 42) and the result is thentransmitted.

Alternatively, phase changing may be performed on both modulated signalss1(t) and s2(t). As such, the transmission device is configured so as toinclude a phase changer taking both signals output by the weighting unit600, as shown in FIG. 26.

Like phase changer 317B, phase changer 317A performs regular a regularchange of phase on the signal input thereto, and as such changes thephase of signal z1′(t) precoded by the weighting unit. Post-phase-changesignal z1(t) is then output to a transmitter.

However, the phase changing rate applied by the phase changers 317A and317B varies simultaneously in order to perform the phase changing shownin FIG. 26. (The following describes a non-limiting example of the phasechanging scheme.) For time u, phase changer 317A from FIG. 26 performsthe change of phase such that z1(t)=y₁(t)z1′(t), while phase changer317B performs the change of phase such that z2(t)=y₂(t)z2′(t). Forexample, as shown in FIG. 26, for time u, y₁(u)=e^(j0) andy₂(u)=e^(−jπ/2), for time u+1, y₁(u+1)=e^(jπ/4) and y₂(u+1)=e^(−j3π/4),and for time u+k, y₁(u+k)=e^(jkπ/4) and y₂(u+k)=e^(j(k3π/4−π/2)). Here,the regular phase changing period (cycle) may be the same for both phasechangers 317A and 317B, or may vary for each.

Also, as described above, a change of phase may be performed beforeprecoding is performed by the weighting unit. In such a case, thetransmission device should be configured as illustrated in FIG. 27.

When a change of phase is carried out on both modulated signals, each ofthe transmit signals is, for example, control information that includesinformation about the phase changing pattern. By obtaining the controlinformation, the reception device knows the phase changing scheme bywhich the transmission device regularly varies the change, i.e., thephase changing pattern, and is thus able to demodulate (decode) thesignals correctly.

Next, variants of the sample configurations shown in FIGS. 6 and 25 aredescribed with reference to FIGS. 28 and 29. FIG. 28 differs from FIG. 6in the inclusion of phase change ON/OFF information 2800 and in that thechange of phase is performed on only one of z1′(t) and z2′(t) (i.e.,performed on one of z1′(t) and z2′(t), which have identical time or acommon frequency). Accordingly, in order to perform the change of phaseon one of z1′(t) and z2′(t), the phase changers 317A and 317B shown inFIG. 28 may each be ON, and performing the change of phase, or OFF, andnot performing the change of phase. The phase change ON/OFF information2800 is control information therefor. The phase change ON/OFFinformation 2800 is output by the signal processing scheme informationgenerator 314 shown in FIG. 3.

Phase changer 317A of FIG. 28 changes the phase to producez1(t)=y₁(t)z1′(t), while phase changer 317B changes the phase to producez2(t)=y₂(t)z2′(t).

Here, a change of phase having a period (cycle) of four is, for example,applied to z1′(t). (Meanwhile, the phase of z2′(t) is not changed.)Accordingly, for time u, y₁(u)=e^(j0) and y₂(u)=1, for time u+1,y₁(u+1)=e^(jπ/2) and y₂(u+1)=1, for time u+2, y₁(u+2)=e^(jπ) andy₂(u+2)=1, and for time u+3, y₁(u+3)=e^(j3π/2) and y₂(u+3)=1.

Next, a change of phase having a period (cycle) of four is, for example,applied to z2′(t). (Meanwhile, the phase of z1′(t) is not changed.)Accordingly, for time u+4, y₁(u+4)=1 and y₂(u+4)=e^(j0), for time u+5,y₁(u+5)=1 and y₂(u+5)=e^(jπ/2) for time u+6, y₁(u+6)=1 andy₂(u+6)=e^(jπ), and for time u+7, y₁(u+7)=1 and y₂(u+7)=e^(j3π/2).

Accordingly, given the above examples.

for any time 8k, y₁(8k)=e^(j0) and y₂(8k)=1,

for any time 8k+1, y₁(8k+1)=e^(jπ/2) and y₂(8k+1)=1,

for any time 8k+2, y₁(8k+2)=e^(jπ) and y₂(8k+2)=1,

for any time 8k+3, y₁(8k+3)=e^(j3π/2) and y₂(8k+3)=1,

for any time 8k+4, y₁(8k+4)=1 and y₂(8k+4)=e^(j0),

for any time 8k+5, y₁(8k+3)=1 and y₂(8k+5)=e^(jπ/2),

for any time 8k+6, y₁(8k+6)=1 and y₂(8k+6)=e^(jπ), and

for any time 8k+7, y₁(8k+7)=1 and y₂(8k+7)=e^(j3π/2).

As described above, there are two intervals, one where the change ofphase is performed on z1′(t) only, and one where the change of phase isperformed on z2′(t) only. Furthermore, the two intervals form a phasechanging period (cycle). While the above explanation describes theinterval where the change of phase is performed on z1′(t) only and theinterval where the change of phase is performed on z2′(t) only as beingequal, no limitation is intended in this manner. The two intervals mayalso differ. In addition, while the above explanation describesperforming a change of phase having a period (cycle) of four on z1′(t)only and then performing a change of phase having a period (cycle) offour on z2′(t) only, no limitation is intended in this manner. Thechanges of phase may be performed on z1′(t) and on z2′(t) in any order(e.g., the change of phase may alternate between being performed onz1′(t) and on z2′(t), or may be performed in random order).

Phase changer 317A of FIG. 29 changes the phase to produces1′(t)=y₁(t)s1(t), while phase changer 317B changes the phase to produces2′(t)=y₂(t)s2(t).

Here, a change of phase having a period (cycle) of four is, for example,applied to s1(t). (Meanwhile, s2(t) remains unchanged). Accordingly, fortime u, y₁(u)=e^(j0) and y₂(u)=1, for time u+1, y₁(u+1)=e^(jπ/2) andy₂(u+1)=1, for time u+2, y₁(u+2)=e^(jπ) and y₂(u+2)=1, and for time u+3,y₁(u+3)=e^(j3π/2) and y₂(u+3)=1.

Next, a change of phase having a period (cycle) of four is, for example,applied to s2(t). (Meanwhile, s1(t) remains unchanged). Accordingly, fortime u+4, y₁(u+4)=1 and y₂(u+4)=e^(j0), for time u+5, y₁(u+5)=1 andy₂(u+5)=e^(jπ/2), for time u+6, y₁(u+6)=1 and y₂(u+6)=e^(jπ), and fortime u+7, y₁(u+7)=1 and y₂(u+7)=e^(j3π/2).

Accordingly, given the above examples,

for any time 8k, y₁(8k)=e^(j0) and y₂(8k)=1,

for any time 8k+1, y₁(8k+1)=e^(jπ/2) and y₂(8k+1)=1,

for any time 8k+2, y₁(8k+2)=e^(jπ) and y₂(8k+2)=1,

for any time 8k+3, y₁(8k+3)=e^(j3π/2) and y₂(8k+3)=1,

for any time 8k+4, y₁(8k+4)=1 and y₂(8k+4)=e^(j0),

for any time 8k+5, y₁(8k+5)=1 and y₂(8k+5)=e^(jπ/2),

for any time 8k+6, y₁(8k+6)=1 and y₂(8k+6)=e^(jπ), and

for any time 8k+7, y₁(8k+7)=1 and y₂(8k+7)=e^(j3π/2).

As described above, there are two intervals, one where the change ofphase is performed on s1(t) only, and one where the change of phase isperformed on s2(t) only. Furthermore, the two intervals form a phasechanging period (cycle). Although the above explanation describes theinterval where the change of phase is performed on s1(t) only and theinterval where the change of phase is performed on s2(t) only as beingequal, no limitation is intended in this manner. The two intervals mayalso differ. In addition, while the above explanation describesperforming the change of phase having a period (cycle) of four on s1(t)only and then performing the change of phase having a period (cycle) offour on s2(t) only, no limitation is intended in this manner. Thechanges of phase may be performed on s1(t) and on s2(t) in any order(e.g., may alternate between being performed on s1(t) and on s2(t), ormay be performed in random order).

Accordingly, the reception conditions under which the reception devicereceives each transmit signal z1(t) and z2(t) are equalized. Byperiodically switching the phase of the symbols in the received signalsz1(t) and z2(t), the ability of the error corrected codes to correcterrors may be improved, thus ameliorating received signal quality in theLOS environment.

Accordingly, Embodiment 2 as described above is able to produce the sameresults as the previously described Embodiment 1.

Although the present Embodiment used a single-carrier scheme, i.e., timedomain phase changing, as an example, no limitation is intended in thisregard. The same effects are also achievable using multi-carriertransmission. Accordingly, the present Embodiment may also be realizedusing, for example, spread-spectrum communications, OFDM, SC-FDMA(Single Carrier Frequency-Division Multiple Access), SC-OFDM, waveletOFDM as described in Non-Patent Literature 7, and so on. As previouslydescribed, while the present Embodiment explains the change of phase aschanging the phase with respect to the time domain t, the phase mayalternatively be changed with respect to the frequency domain asdescribed in Embodiment 1. That is, considering the phase changingscheme in the time domain t described in the present Embodiment andreplacing t with f (f being the ((sub-) carrier) frequency) leads to achange of phase applicable to the frequency domain. Also, as explainedabove for Embodiment 1, the phase changing scheme of the presentEmbodiment is also applicable to changing the phase with respect boththe time domain and the frequency domain.

Accordingly, although FIGS. 6, 25, 26, and 27 illustrate changes ofphase in the time domain, replacing time t with carrier f in each ofFIGS. 6, 25, 26, and 27 corresponds to a change of phase in thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing the change of phase ontime-frequency blocks.

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (preamble, unique word, etc) or symbolstransmitting control information, may be arranged within the frame inany manner.

Embodiment 3

Embodiments 1 and 2, described above, discuss regular changes of phase.Embodiment 3 describes a scheme of allowing the reception device toobtain good received signal quality for data, regardless of thereception device arrangement, by considering the location of thereception device with respect to the transmission device.

Embodiment 3 concerns the symbol arrangement within signals obtainedthrough a change of phase.

FIG. 31 illustrates an example of frame configuration for a portion ofthe symbols within a signal in the time-frequency domain, given atransmission scheme where a regular change of phase is performed for amulti-carrier scheme such as OFDM.

First, an example is explained in which the change of phase is performedone of two baseband signals, precoded as explained in Embodiment 1 (seeFIG. 6).

(Although FIG. 6 illustrates a change of phase in the time domain,switching time t with carrier f in FIG. 6 corresponds to a change ofphase in the frequency domain. In other words, replacing (t) with (t, f)where t is time and f is frequency corresponds to performing phasechanges on time-frequency blocks.)

FIG. 31 illustrates the frame configuration of modulated signal z2′,which is input to phase changer 317B from FIG. 12. Each squarerepresents one symbol (although both signals s1 and s2 are included forprecoding purposes, depending on the precoding matrix, only one ofsignals s1 and s2 may be used).

Consider symbol 3100 at carrier 2 and time $2 of FIG. 31. The carrierhere described may alternatively be termed a sub-carrier.

Within carrier 2, there is a very strong correlation between the channelconditions for symbol 3100 at carrier 2, time $2 and the channelconditions for the time domain nearest-neighbour symbols to time $2,i.e., symbol 3013 at time $1 and symbol 3101 at time $3 within carrier2.

Similarly, for time $2, there is a very strong correlation between thechannel conditions for symbol 3100 at carrier 2, time $2 and the channelconditions for the frequency-domain nearest-neighbour symbols to carrier2, i.e., symbol 3104 at carrier 1, time $2 and symbol 3104 at time $2,carrier 3.

As described above, there is a very strong correlation between thechannel conditions for symbol 3100 and the channel conditions forsymbols 3101, 3102, 3103, and 3104.

The present description considers N different phases (N being aninteger, N≥2) for multiplication in a transmission scheme where thephase is regularly changed. The symbols illustrated in FIG. 31 areindicated as e^(j0), for example. This signifies that this symbol issignal z2′ from FIG. 6 phase-changed through multiplication by e^(j0).That is, the values indicated in FIG. 31 for each of the symbols are thevalues of y(t) from formula 42, which are also the values ofz2(t)=y₂(t)z2′(t) described in Embodiment 2.

The present Embodiment takes advantage of the high correlation inchannel conditions existing between neighboring symbols in the frequencydomain and/or neighbouring symbols in the time domain in a symbolarrangement enabling high data reception quality to be obtained by thereception device receiving the phase-changed symbols.

In order to achieve this high data reception quality, conditions #1 and#2 are necessary.

(Condition #1)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a symbol fortransmitting data (hereinafter, data symbol), neighbouring symbols inthe time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Yare also data symbols, and a different change of phase should beperformed on precoded baseband signal z2′ corresponding to each of thesethree data symbols, i.e., on precoded baseband signal z2′ at time X,carrier Y, at time X−1, carrier Y and at time X+1, carrier Y.

(Condition #2)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a data symbol,neighbouring symbols in the frequency domain, i.e., at time X, carrierY−1 and at time X, carrier Y+1 are also data symbols, and a differentchange of phase should be performed on precoded baseband signal z2′corresponding to each of these three data symbols, i.e., on precodedbaseband signal z2′ at time X, carrier Y, at time X, carrier Y−1 and attime X, carrier Y+1.

Ideally, data symbols satisfying Condition #1 should be present.Similarly, data symbols satisfying Condition #2 should be present.

The reasons supporting Conditions #1 and #2 are as follows.

A very strong correlation exists between the channel conditions of givensymbol of a transmit signal (hereinafter, symbol A) and the channelconditions of the symbols neighbouring symbol A in the time domain, asdescribed above.

Accordingly, when three neighbouring symbols in the time domain eachhave different phases, then despite reception quality degradation in theLOS environment (poor signal quality caused by degradation in conditionsdue to direct wave phase relationships despite high signal quality interms of SNR) for symbol A, the two remaining symbols neighbouringsymbol A are highly likely to provide good reception quality. As aresult, good received signal quality is achievable after errorcorrection and decoding.

Similarly, a very strong correlation exists between the channelconditions of given symbol of a transmit signal (hereinafter, symbol A)and the channel conditions of the symbols neighbouring symbol A in thefrequency domain, as described above.

Accordingly, when three neighbouring symbols in the frequency domaineach have different phases, then despite reception quality degradationin the LOS environment (poor signal quality caused by degradation inconditions due to direct wave phase relationships despite high signalquality in terms of SNR) for symbol A, the two remaining symbolsneighbouring symbol A are highly likely to provide good receptionquality. As a result, good received signal quality is achievable aftererror correction and decoding.

Combining Conditions #1 and #2, ever greater data reception quality islikely achievable for the reception device. Accordingly, the followingCondition #3 can be derived.

(Condition #3)

As shown in FIG. 6, for a transmission scheme involving a regular changeof phase performed on precoded baseband signal z2′ using multi-carriertransmission such as OFDM, time X, carrier Y is a data symbol,neighbouring symbols in the time domain, i.e., at time X−1, carrier Yand at time X+1, carrier Y are also data symbols, and neighbouringsymbols in the frequency domain, i.e., at time X, carrier Y−1 and attime X, carrier Y+1 are also data symbols, and a different change inphase should be performed on precoded baseband signal z2′ correspondingto each of these five data symbols, i.e., on precoded baseband signalz2′ at time X, carrier Y, at time X, carrier Y−1, at time X, carrierY+1, at a time X−1, carrier Y, and at time X+1, carrier Y.

Here, the different changes in phase are as follows. Changes in phaseare defined from 0 radians to 2π radians. For example, for time X,carrier Y, a phase change of e^(jθX,Y) is applied to precoded basebandsignal z2′ from FIG. 6, for time X−1, carrier Y, a phase change ofe^(jθX−1,Y) is applied to precoded baseband signal z2′ from FIG. 6, fortime X+1, carrier Y, a phase change of e^(jθX+1,Y) is applied toprecoded baseband signal z2′ from FIG. 6, such that 0≤θ_(X,Y)<2π,0≤θ_(X−1,Y)<2π, and 0≤_(X+1,Y)<2π, all units being in radians.Accordingly, for Condition #1, it follows that θ_(X,Y)≠θ_(X−1,Y),θ_(X,Y)≠θ_(X+1,Y), and that θ_(X−1,Y)≠θ_(X+1,Y). Similarly, forCondition #2, it follows that θ_(X,Y)≠θ_(X,Y−1), θ_(X,Y)≠θ_(X,Y+1), andthat θ_(X,Y−1)≠θ_(X,Y+1). And, for Condition #3, it follows thatθ_(X,Y)≠θ_(X−1,Y), θ_(X,Y)≠θ_(X+1,Y), θ_(X,Y)≠θ_(X,Y−1),θ_(X,Y)≠θ_(X,Y−1), θ_(X−1,Y)≠θ_(X+1,Y), θ_(X−1,Y)≠θ_(X,Y−1),θ_(X−1,Y)≠θ_(X+1,Y), θ_(X+1,Y)≠θ_(X−1,Y), θ_(X+1,Y)≠θ_(X,Y+1), and thatθ_(X,Y−1)≠θ_(X,Y+1).

Ideally, a data symbol should satisfy Condition #3.

FIG. 31 illustrates an example of Condition #3 where symbol Acorresponds to symbol 3100. The symbols are arranged such that the phaseby which precoded baseband signal z2′ from FIG. 6 is multiplied differsfor symbol 3100, for both neighbouring symbols thereof in the timedomain 3101 and 3102, and for both neighbouring symbols thereof in thefrequency domain 3102 and 3104. Accordingly, despite received signalquality degradation of symbol 3100 for the receiver, good signal qualityis highly likely for the neighbouring signals, thus guaranteeing goodsignal quality after error correction.

FIG. 32 illustrates a symbol arrangement obtained through phase changesunder these conditions.

As evident from FIG. 32, with respect to any data symbol, a differentchange in phase is applied to each neighbouring symbol in the timedomain and in the frequency domain. As such, the ability of thereception device to correct errors may be improved.

In other words, in FIG. 32, when all neighbouring symbols in the timedomain are data symbols, Condition #1 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols, Condition #2 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols and all neighbouring symbols in the time domainare data symbols, Condition #3 is satisfied for all Xs and all Ys.

The following describes an example in which a change of phase isperformed on two precoded baseband signals, as explained in Embodiment 2(see FIG. 26).

When a change of phase is performed on precoded baseband signal z1′ andprecoded baseband signal z2′ as shown in FIG. 26, several phase changingschemes are possible. The details thereof are explained below.

Scheme 1 involves a change in phase performed on precoded basebandsignal z2′ as described above, to achieve the change in phaseillustrated by FIG. 32. In FIG. 32, a change of phase having a period(cycle) of 10 is applied to precoded baseband signal z2′. However, asdescribed above, in order to satisfy Conditions #1, #2, and #3, thechange in phase applied to precoded baseband signal z2′ at each(sub-)carrier varies over time. (Although such changes are applied inFIG. 32 with a period (cycle) of ten, other phase changing schemes arealso possible.) Then, as shown in FIG. 33, the change in phase performedon precoded baseband signal z1′ produces a constant value that isone-tenth of that of the change in phase performed on precoded basebandsignal z2′. In FIG. 33, for a period (cycle) (of change in phaseperformed on precoded baseband signal z2′) including time $1, the valueof the change in phase performed on precoded baseband signal z1′ ise^(j0). Then, for the next period (cycle) (of change in phase performedon precoded baseband signal z2′) including time $2, the value of thechange in phase performed on precoded baseband signal z1′ is e^(jπ/9),and so on.

The symbols illustrated in FIG. 33 are indicated as e^(j0), for example.This signifies that this symbol is signal z1′ from FIG. 26 on which achange in phase as been applied through multiplication by e^(j0). Thatis, the values indicated in FIG. 33 for each of the symbols are thevalues of z1′(t)=y₂(t)z1′(t) described in Embodiment 2 for y₁(t).

As shown in FIG. 33, the change in phase performed on precoded basebandsignal z1′ produces a constant value that is one-tenth that of thechange in phase performed on precoded baseband signal z2′ such that thephase changing value varies with the number of each period (cycle). (Asdescribed above, in FIG. 33, the value is e^(j0) for the first period(cycle), e^(jπ/9) for the second period (cycle), and so on.)

As described above, the change in phase performed on precoded basebandsignal z2′ has a period (cycle) of ten, but the period (cycle) can beeffectively made greater than ten by taking the change in phase appliedto precoded baseband signal z1′ and to precoded baseband signal z2′ intoconsideration. Accordingly, data reception quality may be improved forthe reception device.

Scheme 2 involves a change in phase of precoded baseband signal z2′ asdescribed above, to achieve the change in phase illustrated by FIG. 32.In FIG. 32, a change of phase having a period (cycle) of ten is appliedto precoded baseband signal z2′. However, as described above, in orderto satisfy Conditions #1, #2, and #3, the change in phase applied toprecoded baseband signal z2′ at each (sub-)carrier varies over time.(Although such changes are applied in FIG. 32 with a period (cycle) often, other phase changing schemes are also possible.) Then, as shown inFIG. 30, the change in phase performed on precoded baseband signal z1′differs from that performed on precoded baseband signal z2′ in having aperiod (cycle) of three rather than ten.

The symbols illustrated in FIG. 30 are indicated as e^(j0), for example.This signifies that this symbol is signal z1′ from FIG. 26 to which achange in phase has been applied through multiplication by e^(j0). Thatis, the values indicated in FIG. 30 for each of the symbols are thevalues of z1(t)=y₁(t)z1′(t) described in Embodiment 2 for y₁(t).

As described above, the change in phase performed on precoded basebandsignal z2′ has a period (cycle) of ten, but by taking the changes inphase applied to precoded baseband signal z1′ and precoded basebandsignal z2′ into consideration, the period (cycle) can be effectivelymade equivalent to 30 for both precoded baseband signals z1′ and z2′.Accordingly, data reception quality may be improved for the receptiondevice. An effective way of applying scheme 2 is to perform a change inphase on precoded baseband signal z1′ with a period (cycle) of N andperform a change in phase on precoded baseband signal z2′ with a period(cycle) of M such that N and M are coprime. As such, by taking bothprecoded baseband signals z1′ and z2′ into consideration, a period(cycle) of N×M is easily achievable, effectively making the period(cycle) greater when N and M are coprime.

The above describes an example of the phase changing scheme pertainingto Embodiment 3. The present invention is not limited in this manner. Asexplained for Embodiments 1 and 2, a change in phase may be performedwith respect the frequency domain or the time domain, or ontime-frequency blocks. Similar improvement to the data reception qualitycan be obtained for the reception device in all cases.

The same also applies to frames having a configuration other than thatdescribed above, where pilot symbols (SP (Scattered Pilot) and symbolstransmitting control information are inserted among the data symbols.The details of change in phase in such circumstances are as follows.

FIGS. 47A and 47B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 47A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 47Billustrates the frame configuration of modulated signal (precodedbaseband signals) z2′. In FIGS. 47A and 47B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich precoding or precoding and a change in phase have been performed.

FIGS. 47A and 47B, like FIG. 6, indicate the arrangement of symbols whena change in phase is applied to precoded baseband signal z2′ (while nochange of phase is performed on precoded baseband signal z1). (AlthoughFIG. 6 illustrates a change in phase with respect to the time domain,switching time t with carrier f in FIG. 6 corresponds to a change inphase with respect to the frequency domain. In other words, replacing(t) with (t, f) where t is time and f is frequency corresponds toperforming a change of phase on time-frequency blocks.) Accordingly, thenumerical values indicated in FIGS. 47A and 47B for each of the symbolsare the values of precoded baseband signal z2′ after the change inphase. No values are given for the symbols of precoded baseband signalz1′ (z1) as no change in phase is performed thereon.

The key point of FIGS. 47A and 47B is that the change in phase isperformed on the data symbols of precoded baseband signal z2′, i.e., onprecoded symbols. (The symbols under discussion, being precoded,actually include both symbols s1 and s2.) Accordingly, no change ofphase is performed on the pilot symbols inserted into z2′.

FIGS. 48A and 48B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 48A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 47Billustrates the frame configuration of modulated signal (precodedbaseband signals) z2′. In FIGS. 48A and 48B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich precoding, or precoding and a change in phase, have beenperformed.

FIGS. 48A and 48B, like FIG. 26, indicate the arrangement of symbolswhen a change in phase is applied to precoded baseband signal z1′ and toprecoded baseband signal z2′. (Although FIG. 26 illustrates a change inphase with respect to the time domain, switching time t with carrier fin FIG. 26 corresponds to a change in phase with respect to thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing a change of phase ontime-frequency blocks.) Accordingly, the numerical values indicated inFIGS. 48A and 48B for each of the symbols are the values of precodedbaseband signal z1′ and z2′ after the change in phase.

The key point of FIG. 47 is that a change of phase is performed on thedata symbols of precoded baseband signal z1′, that is, on the precodedsymbols thereof, and on the data symbols of precoded baseband signalz2′, that is, on the precoded symbols thereof. (The symbols underdiscussion, being precoded, actually include both symbols s1 and s2.)Accordingly, no change of phase is performed on the pilot symbolsinserted in z1′, nor on the pilot symbols inserted in z2′.

FIGS. 49A and 49B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 49A illustrates the frame configuration ofmodulated signal (precoded baseband signals) z1 or z1′ while FIG. 49Billustrates the frame configuration of modulated signal (precodedbaseband signal) z2′. In FIGS. 49A and 49B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on which precodingor precoding and the change in phase have been performed. FIGS. 49A and49B differ from FIGS. 47A and 47B in the configuration scheme forsymbols other than data symbols. The times and carriers at which pilotsymbols are inserted into modulated signal z1′ are null symbols inmodulated signal z2′. Conversely, the times and carriers at which pilotsymbols are inserted into modulated signal z2′ are null symbols inmodulated signal z1′.

FIGS. 49A and 49B, like FIG. 6, indicate the arrangement of symbols whena change in phase is applied to precoded baseband signal z2′ (while nochange of phase is performed on precoded baseband signal z1). (AlthoughFIG. 6 illustrates a change of phase with respect to the time domain,switching time t with carrier f in FIG. 6 corresponds to a change ofphase with respect to the frequency domain. In other words, replacing(t) with (t, f) where t is time and f is frequency corresponds toperforming a change of phase on time-frequency blocks.) Accordingly, thenumerical values indicated in FIGS. 49A and 49B for each of the symbolsare the values of precoded baseband signal z2′ after a change of phaseis performed. No values are given for the symbols of precoded basebandsignal z1′ (z1) as no change of phase is performed thereon.

The key point of FIGS. 49A and 49B is that a change of phase isperformed on the data symbols of precoded baseband signal z2′, i.e., onprecoded symbols. (The symbols under discussion, being precoded,actually include both symbols s1 and s2.) Accordingly, no change ofphase is performed on the pilot symbols inserted into z2′.

FIGS. 50A and 50B illustrate the frame configuration of modulatedsignals (precoded baseband signals) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 50A illustrates the frame configuration ofmodulated signal (precoded baseband signal) z1 or z1′ while FIG. 50Billustrates the frame configuration of modulated signal (precodedbaseband signal) z2′. In FIGS. 50A and 50B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on whichprecoding, or precoding and a change of phase, have been performed.FIGS. 50A and 50B differ from FIGS. 48A and 48B in the configurationscheme for symbols other than data symbols. The times and carriers atwhich pilot symbols are inserted into modulated signal z1′ are nullsymbols in modulated signal z2′. Conversely, the times and carriers atwhich pilot symbols are inserted into modulated signal z2′ are nullsymbols in modulated signal z1′.

FIGS. 50A and 50B, like FIG. 26, indicate the arrangement of symbolswhen a change of phase is applied to precoded baseband signal z1′ and toprecoded baseband signal z2′. (Although FIG. 26 illustrates a change ofphase with respect to the time domain, switching time t with carrier fin FIG. 26 corresponds to a change of phase with respect to thefrequency domain. In other words, replacing (t) with (t, f) where t istime and f is frequency corresponds to performing a change of phase ontime-frequency blocks.) Accordingly, the numerical values indicated inFIGS. 50A and 50B for each of the symbols are the values of precodedbaseband signal z1′ and z2′ after a change of phase.

The key point of FIGS. 50A and 50B is that a change of phase isperformed on the data symbols of precoded baseband signal z1′, that is,on the precoded symbols thereof, and on the data symbols of precodedbaseband signal z2′, that is, on the precoded symbols thereof. (Thesymbols under discussion, being precoded, actually include both symbolss1 and s2.) Accordingly, no change of phase is performed on the pilotsymbols inserted in z1′, nor on the pilot symbols inserted in z2′.

FIG. 51 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 47A, 47B, 49A, and 49B. Components thereofperforming the same operations as those of FIG. 4 use the same referencesymbols thereas.

In FIG. 51, the weighting units 308A and 308B and phase changer 317Bonly operate at times indicated by the frame configuration signal 313 ascorresponding to data symbols.

In FIG. 51, a pilot symbol generator 5101 (that also generates nullsymbols) outputs baseband signals 5102A and 5102B for a pilot symbolwhenever the frame configuration signal 313 indicates a pilot symbol (ora null symbol).

Although not indicated in the frame configurations from FIGS. 47Athrough 50B, when precoding (or phase rotation) is not performed, suchas when transmitting a modulated signal using only one antenna (suchthat the other antenna transmits no signal) or when using a space-timecoding transmission scheme (particularly, space-time block coding) totransmit control information symbols, then the frame configurationsignal 313 takes control information symbols 5104 and controlinformation 5103 as input. When the frame configuration signal 313indicates a control information symbol, baseband signals 5102A and 5102Bthereof are output.

Wireless units 310A and 310B of FIG. 51 take a plurality of basebandsignals as input and select a desired baseband signal according to theframe configuration signal 313. Wireless units 310A and 310B then applyOFDM signal processing and output modulated signals 311A and 311Bconforming to the frame configuration.

FIG. 52 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 48A, 48B, 50A, and 50B. Components thereofperforming the same operations as those of FIGS. 4 and 51 use the samereference symbols thereas. FIG. 51 features an additional phase changer317A that only operates when the frame configuration signal 313indicates a data symbol. At all other times, the operations areidentical to those explained for FIG. 51.

FIG. 53 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 51. The following describes the points ofdifference. As shown in FIG. 53, phase changer 317B takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 317B performs a change of phaseon precoded baseband signal 316B. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations, such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

A selector 5301 takes the plurality of baseband signals as input andselects a baseband signal having a symbol indicated by the frameconfiguration signal 313 for output.

FIG. 54 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 52. The following describes the points ofdifference. As shown in FIG. 54, phase changer 317B takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 317B performs a change of phaseon precoded baseband signal 316B. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

Similarly, as shown in FIG. 54, phase changer 5201 takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 5201 performs a change of phaseon precoded baseband signal 309A. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 5201 pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

The above explanations are given using pilot symbols, control symbols,and data symbols as examples. However, the present invention is notlimited in this manner. When symbols are transmitted using schemes otherthan precoding, such as single-antenna transmission or transmissionusing space-time block coding, not performing a change of phase isimportant. Conversely, performing a change of phase on symbols that havebeen precoded is the key point of the present invention.

Accordingly, a characteristic feature of the present invention is thatthe change of phase is not performed on all symbols within the frameconfiguration in the time-frequency domain, but only performed onsignals that have been precoded.

Embodiment 4

Embodiments 1 and 2, described above, discuss a regular change of phase.Embodiment 3, however, discloses performing a different change of phaseon neighbouring symbols.

The present Embodiment describes a phase changing scheme that variesaccording to the modulation scheme and the coding rate of theerror-correcting codes used by the transmission device.

Table 1, below, is a list of phase changing scheme settingscorresponding to the settings and parameters of the transmission device.

TABLE 1 No. of Modulated Phase Transmission Changing Signals ModulationScheme Coding Rate Pattern 2 #1: QPSK, #2: QPSK #1: 1/2, #2 2/3 #1: —,#2: A 2 #1: QPSK, #2: QPSK #1: 1/2, #2: 3/4 #1: A, #2: B 2 #1: QPSK, #2:QPSK #1: 2/3, #2: 3/5 #1: A, #2: C 2 #1: QPSK, #2: QPSK #1: 2/3, #2: 2/3#1: C, #2: — 2 #1: QPSK, #2: QPSK #1: 3/3, #2: 2/3 #1: D, #2: E 2 #1:QPSK, #2: 16-QAM #1: 1/2, #2: 2/3 #1: B, #2: A 2 #1: QPSK, #2: 16-QAM#1: 1/2, #2: 3/4 #1: A, #2: C 2 #1: QPSK, #2: 16-QAM #1: 1/2, #2: 3/5#1: —, #2: E 2 #1: QPSK, #2: 16-QAM #1: 2/3, #2: 3/4 #1: D, #2: — 2 #1:QPSK, #2: 16-QAM #1: 2/3, #2: 5/6 #1: D, #2: B 2 #1: 16-QAM, #2: 16-QAM#1: 1/2, #2: 2/3 #1: —, #2: E . . . . . . . . . . . .

In Table 1, #1 denotes modulated signal s1 from Embodiment 1 describedabove (baseband signal s1 modulated with the modulation scheme set bythe transmission device) and #2 denotes modulated signal s2 (basebandsignal s2 modulated with the modulation scheme set by the transmissiondevice). The coding rate column of Table 1 indicates the coding rate ofthe error-correcting codes for modulation schemes #1 and #2. The phasechanging pattern column of Table 1 indicates the phase changing schemeapplied to precoded baseband signals z1 (z1′) and z2 (z2′), as explainedin Embodiments 1 through 3. Although the phase changing patterns arelabeled A, B, C, D, E, and so on, this refers to the phase change degreeapplied, for example, in a phase changing pattern given by formula 46and formula 47, above. In the phase changing pattern column of Table 1,the dash signifies that no change of phase is applied.

The combinations of modulation scheme and coding rate listed in Table 1are examples. Other modulation schemes (such as 128-QAM and 256-QAM) andcoding rates (such as 7/8) not listed in Table 1 may also be included.Also, as described in Embodiment 1, the error-correcting codes used fors1 and s2 may differ (Table 1 is given for cases where a single type oferror-correcting codes is used, as in FIG. 4). Furthermore, the samemodulation scheme and coding rate may be used with different phasechanging patterns. The transmission device transmits informationindicating the phase changing patterns to the reception device. Thereception device specifies the phase changing pattern bycross-referencing the information and Table 1, then performsdemodulation and decoding. When the modulation scheme anderror-correction scheme determine a unique phase changing pattern, thenas long as the transmission device transmits the modulation scheme andinformation regarding the error-correction scheme, the reception deviceknows the phase changing pattern by obtaining that information. As such,information pertaining to the phase changing pattern is not strictlynecessary.

In Embodiments 1 through 3, the change of phase is applied to precodedbaseband signals. However, the amplitude may also be modified along withthe phase in order to apply periodical, regular changes. Accordingly, anamplification modification pattern regularly modifying the amplitude ofthe modulated signals may also be made to conform to Table 1. In suchcircumstances, the transmission device should include an amplificationmodifier that modifies the amplification after weighting unit 308A orweighting unit 308B from FIG. 3 or 4. In addition, amplificationmodification may be performed on only one of or on both of the precodedbaseband signals z1(t) and z2(t) (in the former case, the amplificationmodifier is only needed after one of weighting unit 308A and 308B).

Furthermore, although not indicated in Table 1 above, the mapping schememay also be regularly modified by the mapper, without a regular changeof phase.

That is, when the mapping scheme for modulated signal s1(t) is 16-QAMand the mapping scheme for modulated signal s2(t) is also 16-QAM, themapping scheme applied to modulated signal s2(t) may be regularlychanged as follows: from 16-QAM to 16-APSK, to 16-QAM in the IQ plane,to a first mapping scheme producing a signal point layout unlike16-APSK, to 16-QAM in the IQ plane, to a second mapping scheme producinga signal point layout unlike 16-APSK, and so on. As such, the datareception quality can be improved for the reception device, much likethe results obtained by a regular change of phase described above.

In addition, the present invention may use any combination of schemesfor a regular change of phase, mapping scheme, and amplitude, and thetransmit signal may transmit with all of these taken into consideration.

The present Embodiment may be realized using single-carrier schemes aswell as multi-carrier schemes. Accordingly, the present Embodiment mayalso be realized using, for example, spread-spectrum communications,OFDM, SC-FDMA, SC-OFDM, wavelet OFDM as described in Non-PatentLiterature 7, and so on. As described above, the present Embodimentdescribes changing the phase, amplitude, and mapping schemes byperforming phase, amplitude, and mapping scheme modifications withrespect to the time domain t. However, much like Embodiment 1, the samechanges may be carried out with respect to the frequency domain. Thatis, considering the phase, amplitude, and mapping scheme modification inthe time domain t described in the present Embodiment and replacing twith f (f being the ((sub-) carrier) frequency) leads to phase,amplitude, and mapping scheme modification applicable to the frequencydomain. Also, the phase, amplitude, and mapping scheme modification ofthe present Embodiment is also applicable to phase, amplitude, andmapping scheme modification in both the time domain and the frequencydomain.

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (preamble, unique word, etc) or symbolstransmitting control information, may be arranged within the frame inany manner.

Embodiment A1

The present Embodiment describes a scheme for regularly changing thephase when encoding is performed using block codes as described inNon-Patent Literature 12 through 15, such as QC (Quasi-Cyclic) LDPCCodes (not only QC-LDPC but also LDPC codes may be used), concatenatedLDPC and BCH (Bose-Chaudhuri-Hocquenghem) codes, Turbo codes orDuo-Binary Turbo Codes using tail-biting, and so on. The followingexample considers a case where two streams s1 and s2 are transmitted.However, when encoding has been performed using block codes and controlinformation and the like is not required, the number of bits making upeach coded block matches the number of bits making up each block code(control information and so on described below may yet be included).When encoding has been performed using block codes or the like andcontrol information or the like (e.g., CRC (cyclic redundancy check)transmission parameters) is required, then the number of bits making upeach coded block is the sum of the number of bits making up the blockcodes and the number of bits making up the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed ineach coded block when block codes are used. FIG. 34 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 4, and thetransmission device has only one encoder. (Here, the transmission schememay be any single-carrier scheme or multi-carrier scheme such as OFDM.)

As shown in FIG. 34, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

Then, given that the transmission device from FIG. 4 transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols (hereinafter, slots) are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up a single coded block,and when the modulation scheme is 64-QAM, 500 slots are needed totransmit all of the bits making up a single coded block.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase. That is, five different phase changing values (or phasechanging sets) have been prepared for the phase changer of thetransmission device from FIG. 4 (equivalent to the period (cycle) fromEmbodiments 1 through 4) (As in FIG. 6, five phase changing values areneeded in order to perform a change of phase with a period (cycle) offive on precoded baseband signal z2′ only. Also, as in FIG. 26, twophase changing values are needed for each slot in order to perform thechange of phase on both precoded baseband signals z1′ and z2′. These twophase changing values are termed a phase changing set. Accordingly, fivephase changing sets should ideally be prepared in order to perform thechange of phase with a period (cycle) of five in such circumstances).These five phase changing values (or phase changing sets) are expressedas PHASE[0], PHASE[1], PHASE[2], PHASE[3], and PHASE[4].

For the above-described 1500 slots needed to transmit the 6000 bitsmaking up a single coded block when the modulation scheme is QPSK,PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2]is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] isused on 300 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Similarly, for the above-described 700 slots needed to transmit the 6000bits making up a single coded block when the modulation scheme is16-QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 150 slots,PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, andPHASE[4] is used on 150 slots.

Furthermore, for the above-described 500 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is64-QAM, PHASE[0] is used on 100 slots, PHASE[1] is used on 100 slots,PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, andPHASE[4] is used on 100 slots.

As described above, a scheme for a regular change of phase requires thepreparation of N phase changing values (or phase changing sets) (wherethe N different phases are expressed as PHASE[0], PHASE[1], PHASE[2] . .. PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bitsmaking up a single coded block, PHASE[0] is used on K₀ slots, PHASE[1]is used on K₁ slots, PHASE[i] is used on K_(i) slots (where i=0, 1, 2 .. . N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1]is used on K_(N−1) slots, such that Condition #A01 is met.

(Condition #A01)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≤a≤N−1, bdenotes an integer that satisfies 0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported modulation scheme for use, Condition#A01 is preferably satisfied for the supported modulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #A01 may not be satisfied for some modulation schemes. In sucha case, the following condition applies instead of Condition #A01.

(Condition #A02)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

FIG. 35 illustrates the varying numbers of symbols and slots needed intwo coded blocks when block codes are used. FIG. 35 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 3 and FIG.12, and the transmission device has two encoders. (Here, thetransmission scheme may be any single-carrier scheme or multi-carrierscheme such as OFDM.)

As shown in FIG. 35, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

The transmission device from FIG. 3 and the transmission device fromFIG. 12 each transmit two streams at once, and have two encoders. Assuch, the two streams each transmit different code blocks. Accordingly,when the modulation scheme is QPSK, two coded blocks drawn from s1 ands2 are transmitted within the same interval, e.g., a first coded blockdrawn from s1 is transmitted, then a second coded block drawn from s2 istransmitted. As such, 3000 slots are needed in order to transmit thefirst and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 slotsare needed to transmit all of the bits making up the two coded blocks,and when the modulation scheme is 64-QAM, 1000 slots are needed totransmit all of the bits making up the two coded blocks.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase. That is, five different phase changing values (or phasechanging sets) have been prepared for the phase changers of thetransmission devices from FIGS. 3 and 12 (equivalent to the period(cycle) from Embodiments 1 through 4) (As in FIG. 6, five phase changingvalues are needed in order to perform a change of phase having a period(cycle) of five on precoded baseband signal z2′ only. Also, as in FIG.26, two phase changing values are needed for each slot in order toperform the change of phase on both precoded baseband signals z1′ andz2′. These two phase changing values are termed a phase changing set.Accordingly, five phase changing sets should ideally be prepared inorder to perform the change of phase with a period (cycle) of five insuch circumstances). These five phase changing values (or phase changingsets) are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[3], andPHASE[4].

For the above-described 3000 slots needed to transmit the 6000×2 bitsmaking up a single coded block when the modulation scheme is QPSK,PHASE[0] is used on 600 slots, PHASE[1] is used on 600 slots, PHASE[2]is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] isused on 600 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2]is used on slots 600 times, PHASE[3] is used on slots 600 times, andPHASE[4] is used on slots 600 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 600 times, PHASE[1] isused on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3]is used on slots 600 times, and PHASE[4] is used on slots 600 times.

Similarly, for the above-described 1500 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is16-QAM, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots,PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, andPHASE[4] is used on 300 slots.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2]is used on slots 300 times, PHASE[3] is used on slots 300 times, andPHASE[4] is used on slots 300 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 300 times, PHASE[1] isused on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3]is used on slots 300 times, and PHASE[4] is used on slots 300 times.

Similarly, for the above-described 1000 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is64-QAM, PHASE[0] is used on 200 slots, PHASE[1] is used on 200 slots,PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, andPHASE[4] is used on 200 slots.

Furthermore, in order to transmit the first coded block, PHASE[0] isused on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2]is used on slots 200 times, PHASE[3] is used on slots 200 times, andPHASE[4] is used on slots 200 times. Furthermore, in order to transmitthe second coded block, PHASE[0] is used on slots 200 times, PHASE[1] isused on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3]is used on slots 200 times, and PHASE[4] is used on slots 200 times.

As described above, a scheme for regularly changing the phase requiresthe preparation of phase changing values (or phase changing sets)expressed as PHASE[0], PHASE[1], PHASE[2] . . . PHASE[N−2], PHASE[N−1].As such, in order to transmit all of the bits making up two codedblocks, PHASE[0] is used on K₀ slots, PHASE[1] is used on K₁ slots,PHASE[i] is used on K_(i) slots (where i=0, 1, 2 . . . N−1 (i denotes aninteger that satisfies 0≤i≤N−1), and PHASE[N−1] is used on K_(N−1)slots, such that Condition #A03 is met.

(Condition #A03)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≤a≤N−1, bdenotes an integer that satisfies 0≤b≤N−1), a≠b).

Further, in order to transmit all of the bits making up the first codedblock, PHASE[0] is used K_(0,1) times, PHASE[1] is used K_(1,1) times,PHASE[i] is used K_(i,1) times (where i=0, 1, 2 . . . N−1(i denotes aninteger that satisfies 0≤i≤N−1), and PHASE[N−1] is used K_(N−1,1) times,such that Condition #A04 is met.

(Condition #A04)

K_(0,1)=K_(1,1)= . . . K_(i,1)= . . . K_(N−1,1). That is,K_(a,1)=K_(b,1) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

Furthermore, in order to transmit all of the bits making up the secondcoded block, PHASE[0] is used K_(0,2) times, PHASE[1] is used K_(1,2)times, PHASE[i] is used K_(i,2) times (where i=0, 1, 2 . . . N−1 (idenotes an integer that satisfies 0≤i≤N−1), and PHASE[N−1] is usedK_(N−1,2) times, such that Condition #A05 is met.

(Condition #A05)

K_(0,2)=K_(1,2)= . . . K_(i,2)= . . . K_(N−1,2). That is,K_(a,2)=K_(b,2) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported modulation scheme for use, Condition#A03, #A04, and #A05 should preferably be met for the supportedmodulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbol (though some may happen to use the same number),Conditions #A03, #A04, and #A05 may not be satisfied for some modulationschemes. In such a case, the following conditions apply instead ofCondition #A03, #A04, and #A05.

(Condition #A06)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

(Condition #A07)

The difference between K_(a,1) and K_(b,1) satisfies 0 or 1. That is,|K_(a,1)−K_(b,1) satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1,(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1) a≠b)

(Condition #A08)

The difference between K_(a,2) and K_(b,2) satisfies 0 or 1. That is,|K_(a,2)−K_(b,2)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

As described above, bias among the phases being used to transmit thecoded blocks is removed by creating a relationship between the codedblock and the phase of multiplication. As such, data reception qualitycan be improved for the reception device.

In the present Embodiment N phase changing values (or phase changingsets) are needed in order to perform a change of phase having a period(cycle) of N with the scheme for a regular change of phase. As such, Nphase changing values (or phase changing sets) PHASE[0], PHASE[1],PHASE[2] . . . PHASE[N−2], and PHASE[N−1] are prepared. However, schemesexist for reordering the phases in the stated order with respect to thefrequency domain. No limitation is intended in this regard. The N phasechanging values (or phase changing sets) may also change the phases ofblocks in the time domain or in the time-frequency domain to obtain asymbol arrangement as described in Embodiment 1. Although the aboveexamples discuss a phase changing scheme with a period (cycle) of N, thesame effects are obtainable using N phase changing values (or phasechanging sets) at random. That is, the N phase changing values (or phasechanging sets) need not always for a regular period (cycle). As long asthe above-described conditions are satisfied, great quality datareception improvements are realizable for the reception device.

Furthermore, given the existence of modes for spatial multiplexing MIMOschemes, MIMO schemes using a fixed precoding matrix, space-time blockcoding schemes, single-stream transmission, and schemes using a regularchange of phase (the transmission schemes described in Embodiments 1through 4), the transmission device (broadcaster, base station) mayselect any one of these transmission schemes.

As described in Non-Patent Literature 3, spatial multiplexing MIMOschemes involve transmitting signals s1 and s2, which are mapped using aselected modulation scheme, on each of two different antennas. Asdescribed in Embodiments 1 through 4, MIMO schemes using a fixedprecoding matrix involve performing precoding only (with no change ofphase). Further, space-time block coding schemes are described inNon-Patent Literature 9, 16, and 17. Single-stream transmission schemesinvolve transmitting signal s1, mapped with a selected modulationscheme, from an antenna after performing predetermined processing.

Schemes using multi-carrier transmission such as OFDM involve a firstcarrier group made up of a plurality of carriers and a second carriergroup made up of a plurality of carriers different from the firstcarrier group, and so on, such that multi-carrier transmission isrealized with a plurality of carrier groups. For each carrier group, anyof spatial multiplexing MIMO schemes, MIMO schemes using a fixedprecoding matrix, space-time block coding schemes, single-streamtransmission, and schemes using a regular change of phase may be used.In particular, schemes using a regular change of phase on a selected(sub-)carrier group are preferably used to realize the presentEmbodiment.

When a change of phase is performed, then for example, a phase changingvalue for PHASE[i] of X radians is performed on only one precodedbaseband signal, the phase changers of FIGS. 3, 4, 5, 12, 25, 29, 51,and 53 multiplies precoded baseband signal z2′ by e^(jx). Then, for achange of phase by, for example, a phase changing set for PHASE[i] of Xradians and Y radians is performed on both precoded baseband signals,the phase changers from FIGS. 26, 27, 28, 52, and 54 multiplies precodedbaseband signal z2′ by e^(jX) and multiplies precoded baseband signalz1′ by e^(jY).

Embodiment B1

The following describes a sample configuration of an application of thetransmission schemes and reception schemes discussed in the aboveembodiments and a system using the application.

FIG. 36 illustrates the configuration of a system that includes devicesexecuting transmission schemes and reception schemes described in theabove Embodiments. As shown in FIG. 36, the devices executingtransmission schemes and reception schemes described in the aboveEmbodiments include various receivers such as a broadcaster, atelevision 3611, a DVD recorder 3612, a STB (set-top box) 3613, acomputer 3620, a vehicle-mounted television 3641, a mobile phone 3630and so on within a digital broadcasting system 3600. Specifically, thebroadcaster 3601 uses a transmission scheme discussed in theabove-described Embodiments to transmit multiplexed data, in whichvideo, audio, and other data are multiplexed, over a predeterminedtransmission band.

The signals transmitted by the broadcaster 3601 are received by anantenna (such as antenna 3660 or 3640) embedded within or externallyconnected to each of the receivers. Each receiver obtains themultiplexed data by using reception schemes discussed in theabove-described Embodiments to demodulate the signals received by theantenna. Accordingly, the digital broadcasting system 3600 is able torealize the effects of the present invention, as discussed in theabove-described Embodiments.

The video data included in the multiplexed data are coded with a videocoding method compliant with a standard such as MPEG-2 (Moving PictureExperts Group), MPEG4-AVC (Advanced Video Coding), VC-1, or the like.The audio data included in the multiplexed data are encoded with anaudio coding method compliant with a standard such as Dolby AC-3 (AudioCoding), Dolby Digital Plus, MLP (Meridian Lossless Packing), DTS(Digital Theater Systems), DTS-HD, PCM (Pulse-Code Modulation), or thelike.

FIG. 37 illustrates the configuration of a receiver 7900 that executes areception scheme described in the above-described Embodiments. Thereceiver 3700 corresponds to a receiver included in one of thetelevision 3611, the DVD recorder 3612, the STB 3613, the computer 3620,the vehicle-mounted television 3641, the mobile phone 3630 and so onfrom FIG. 36. The receiver 3700 includes a tuner 3701 converting ahigh-frequency signal received by an antenna 3760 into a basebandsignal, and a demodulator 3702 demodulating the baseband signal soconverted to obtain the multiplexed data. The demodulator 3702 executesa reception scheme discussed in the above-described Embodiments, andthus achieves the effects of the present invention as explained above.

The receiver 3700 further includes a stream interface 3720 thatdemultiplexes the audio and video data in the multiplexed data obtainedby the demodulator 3702, a signal processor 3704 that decodes the videodata obtained from the demultiplexed video data into a video signal byapplying a video decoding method corresponding thereto and decodes theaudio data obtained from the demultiplexed audio data into an audiosignal by applying an audio decoding method corresponding thereto, anaudio output unit 3706 that outputs the decoded audio signal through aspeaker or the like, and a video display unit 3707 that outputs thedecoded video signal on a display or the like.

When, for example, a user uses a remote control 3750, information for aselected channel (selected (television) program or audio broadcast) istransmitted to an operation input unit 3710. Then, the receiver 3700performs processing on the received signal received by the antenna 3760that includes demodulating the signal corresponding to the selectedchannel, performing error-correcting decoding, and so on, in order toobtain the received data. At this point, the receiver 3700 obtainscontrol symbol information that includes information on the transmissionscheme (the transmission scheme, modulation scheme, error-correctionscheme, and so on from the above-described Embodiments) (as describedusing FIGS. 5 and 41) from control symbols included the signalcorresponding to the selected channel. As such, the receiver 3700 isable to correctly set the reception operations, demodulation scheme,error-correction scheme and so on, thus enabling the data included inthe data symbols transmitted by the broadcaster (base station) to beobtained. Although the above description is given for an example of theuser using the remote control 3750, the same operations apply when theuser presses a selection key embedded in the receiver 3700 to select achannel.

According to this configuration, the user is able to view programsreceived by the receiver 3700.

The receiver 3700 pertaining to the present Embodiment further includesa drive 3708 that may be a magnetic disk, an optical disc, anon-volatile semiconductor memory, or a similar recording medium. Thereceiver 3700 stores data included in the demultiplexed data obtainedthrough demodulation by the demodulator 3702 and error-correctingdecoding (in some circumstances, the data obtained through demodulationby the demodulator 3702 may not be subject to error correction. Also,the receiver 3700 may perform further processing after error correction.The same hereinafter applies to similar statements concerning othercomponents), data corresponding to such data (e.g., data obtainedthrough compression of such data), data obtained through audio and videoprocessing, and so on, on the drive 3708. Here, an optical disc is arecording medium, such as DVD (Digital Versatile Disc) or BD (Blu-rayDisc), that is readable and writable with the use of a laser beam. Amagnetic disk is a floppy disk, a hard disk, or similar recording mediumon which information is storable through the use of magnetic flux tomagnetize a magnetic body. A non-volatile semiconductor memory is arecording medium, such as flash memory or ferroelectric random accessmemory, composed of semiconductor element(s). Specific examples ofnon-volatile semiconductor memory include an SD card using flash memoryand a Flash SSD (Solid State Drive). Naturally, the specific types ofrecording media mentioned herein are merely examples. Other types ofrecording mediums may also be used.

According to this structure, the user is able to record and storeprograms received by the receiver 3700, and is thereby able to viewprograms at any given time after broadcasting by reading out therecorded data thereof.

Although the above explanations describe the receiver 3700 storingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding on the drive 3708, a portion of the dataincluded in the multiplexed data may instead be extracted and recorded.For example, when data broadcasting services or similar content isincluded along with the audio and video data in the multiplexed dataobtained through demodulation by the demodulator 3702 anderror-correcting decoding, the audio and video data may be extractedfrom the multiplexed data demodulated by the demodulator 3702 and storedas new multiplexed data. Furthermore, the drive 3708 may store eitherthe audio data or the video data included in the multiplexed dataobtained through demodulation by the demodulator 3702 anderror-correcting decoding as new multiplexed data. The aforementioneddata broadcasting service content included in the multiplexed data mayalso be stored on the drive 3708.

Furthermore, when a television, recording device (e.g., a DVD recorder,BD recorder HDD recorder, SD card, or similar), or mobile phoneincorporating the receiver 3700 of the present invention receivesmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding that includes data for correcting bugs insoftware used to operate the television or recording device, forcorrecting bugs in software for preventing personal information andrecorded data from being leaked, and so on, such software bugs may becorrected by installing the data on the television or recording device.As such, bugs in the receiver 3700 are corrected through the inclusionof data for correcting bugs in the software of the receiver 3700.Accordingly, the television, recording device, or mobile phoneincorporating the receiver 3700 may be made to operate more reliably.

Here, the process of extracting a portion of the data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is performed by, for example, the streaminterface 3703. Specifically, the stream interface 3703, demultiplexesthe various data included in the multiplexed data demodulated by thedemodulator 3702, such as audio data, video data, data broadcastingservice content, and so on, as instructed by a non-diagrammed controllersuch as a CPU. The stream interface 3703 then extracts and multiplexesonly the indicated demultiplexed data, thus generating new multiplexeddata. The data to be extracted from the demultiplexed data may bedetermined by the user or may be determined in advance according to thetype of recording medium.

According to such a structure, the receiver 3700 is able to extract andrecord only the data needed in order to view the recorded program. Assuch, the amount of data to be recorded can be reduced.

Although the above explanation describes the drive 3708 as storingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, the video data included in themultiplexed data so obtained may be converted by using a different videocoding method than the original video coding method applied thereto, soas to reduce the amount of data or the bit rate thereof. The drive 3708may then store the converted video data as new multiplexed data. Here,the video coding method used to generate the new video data may conformto a different standard than that used to generate the original videodata. Alternatively, the same video coding method may be used withdifferent parameters. Similarly, the audio data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding may be converted by using a differentaudio coding method than the original audio coding method appliedthereto, so as to reduce the amount of data or the bit rate thereof. Thedrive 3708 may then store the converted audio data as new multiplexeddata.

Here, the process by which the audio or video data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is converted so as to reduce the amount ofdata or the bit rate thereof is performed by, for example, the streaminterface 3703 or the signal processor 3704. Specifically, the streaminterface 3703 demultiplexes the various data included in themultiplexed data demodulated by the demodulator 3702, such as audiodata, video data, data broadcasting service content, and so on, asinstructed by an undiagrammed controller such as a CPU. The signalprocessor 3704 then performs processing to convert the video data sodemultiplexed by using a different video coding method than the originalvideo coding method applied thereto, and performs processing to convertthe audio data so demultiplexed by using a different video coding methodthan the original audio coding method applied thereto. As instructed bythe controller, the stream interface 3703 then multiplexes the convertedaudio and video data, thus generating new multiplexed data. The signalprocessor 3704 may, in accordance with instructions from the controller,performing conversion processing on either the video data or the audiodata, alone, or may perform conversion processing on both types of data.In addition, the amounts of video data and audio data or the bit ratethereof to be obtained by conversion may be specified by the user ordetermined in advance according to the type of recording medium.

According to such a structure, the receiver 3700 is able to modify theamount of data or the bitrate of the audio and video data for storageaccording to the data storage capacity of the recording medium, oraccording to the data reading or writing speed of the drive 3708.Therefore, programs can be stored on the drive despite the storagecapacity of the recording medium being less than the amount ofmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, or the data reading or writing speed ofthe drive being lower than the bit rate of the demultiplexed dataobtained through demodulation by the demodulator 3702. As such, the useris able to view programs at any given time after broadcasting by readingout the recorded data.

The receiver 3700 further includes a stream output interface 3709 thattransmits the multiplexed data demultiplexed by the demodulator 3702 toexternal devices through a communications medium 3730. The stream outputinterface 3709 may be, for example, a wireless communication devicetransmitting modulated multiplexed data to an external device using awireless transmission scheme conforming to a wireless communicationstandard such as Wi-Fi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE802.11n, and so on), WiGig, WirelessHD, Bluetooth, ZigBee, and so onthrough a wireless medium (corresponding to the communications medium3730). The stream output interface 3709 may also be a wiredcommunication device transmitting modulated multiplexed data to anexternal device using a communication scheme conforming to a wiredcommunication standard such as Ethernet™, USB (Universal Serial Bus),PLC (Power Line Communication), HDMI (High-Definition MultimediaInterface) and so on through a wired transmission path (corresponding tothe communications medium 3730) connected to the stream output interface3709.

According to this configuration, the user is able to use an externaldevice with the multiplexed data received by the receiver 3700 using thereception scheme described in the above-described Embodiments. The usageof multiplexed data by the user here includes use of the multiplexeddata for real-time viewing on an external device, recording of themultiplexed data by a recording unit included in an external device, andtransmission of the multiplexed data from an external device to a yetanother external device.

Although the above explanations describe the receiver 3700 outputtingmultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding through the stream output interface 3709,a portion of the data included in the multiplexed data may instead beextracted and output. For example, when data broadcasting services orsimilar content is included along with the audio and video data in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding, the audio and video data may be extractedfrom the multiplexed data obtained through demodulation by thedemodulator 3702 and error-correcting decoding, multiplexed and outputby the stream output interface 3709 as new multiplexed data. Inaddition, the stream output interface 3709 may store either the audiodata or the video data included in the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding asnew multiplexed data.

Here, the process of extracting a portion of the data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is performed by, for example, the streaminterface 3703. Specifically, the stream interface 3703 demultiplexesthe various data included in the multiplexed data demodulated by thedemodulator 3702, such as audio data, video data, data broadcastingservice content, and so on, as instructed by an undiagrammed controllersuch as a CPU. The stream interface 3703 then extracts and multiplexesonly the indicated demultiplexed data, thus generating new multiplexeddata. The data to be extracted from the demultiplexed data may bedetermined by the user or may be determined in advance according to thetype of stream output interface 3709.

According to this structure, the receiver 3700 is able to extract andoutput only the required data to an external device. As such, fewermultiplexed data are output using less communication bandwidth.

Although the above explanation describes the stream output interface3709 as outputting multiplexed data obtained through demodulation by thedemodulator 3702 and error-correcting decoding, the video data includedin the multiplexed data so obtained may be converted by using adifferent video coding method than the original video coding methodapplied thereto, so as to reduce the amount of data or the bit ratethereof. The stream output interface 3709 may then output the convertedvideo data as new multiplexed data. Here, the video coding method usedto generate the new video data may conform to a different standard thanthat used to generate the original video data. Alternatively, the samevideo coding method may be used with different parameters. Similarly,the audio data included in the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding maybe converted by using a different audio coding method than the originalaudio coding method applied thereto, so as to reduce the amount of dataor the bit rate thereof. The stream output interface 3709 may thenoutput the converted audio data as new multiplexed data.

Here, the process by which the audio or video data included in themultiplexed data obtained through demodulation by the demodulator 3702and error-correcting decoding is converted so as to reduce the amount ofdata or the bit rate thereof is performed by, for example, the streaminterface 3703 or the signal processor 3704. Specifically, the streaminterface 3703 demultiplexes the various data included in themultiplexed data demodulated by the demodulator 3702, such as audiodata, video data, data broadcasting service content, and so on, asinstructed by an undiagrammed controller. The signal processor 3704 thenperforms processing to convert the video data so demultiplexed by usinga different video coding method than the original video coding methodapplied thereto, and performs processing to convert the audio data sodemultiplexed by using a different video coding method than the originalaudio coding method applied thereto. As instructed by the controller,the stream interface 3703 then multiplexes the converted audio and videodata, thus generating new multiplexed data. The signal processor 3704may, in accordance with instructions from the controller, performingconversion processing on either the video data or the audio data, alone,or may perform conversion processing on both types of data. In addition,the amounts of video data and audio data or the bit rate thereof to beobtained by conversion may be specified by the user or determined inadvance according to the type of stream output interface 3709.

According to this structure, the receiver 3700 is able to modify the bitrate of the video and audio data for output according to the speed ofcommunication with the external device. Thus, despite the speed ofcommunication with an external device being slower than the bit rate ofthe multiplexed data obtained through demodulation by the demodulator3702 and error-correcting decoding, by outputting new multiplexed datafrom the stream output interface to the external device, the user isable to use the new multiplexed data with other communication devices.

The receiver 3700 further includes an audiovisual output interface 3711that outputs audio and video signals decoded by the signal processor3704 to the external device through an external communications medium.The audiovisual output interface 3711 may be, for example, a wirelesscommunication device transmitting modulated audiovisual data to anexternal device using a wireless transmission scheme conforming to awireless communication standard such as Wi-Fi™ (IEEE 802.11a, IEEE802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD,Bluetooth, ZigBee, and so on through a wireless medium. The streamoutput interface 3709 may also be a wired communication devicetransmitting modulated audiovisual data to an external device using acommunication scheme conforming to a wired communication standard suchas Ethernet™, USB, PLC, HDMI, and so on through a wired transmissionpath connected to the stream output interface 3709. Furthermore, thestream output interface 3709 may be a terminal for connecting a cablethat outputs analogue audio signals and video signals as-is.

According to such a structure, the user is able to use the audio signalsand video signals decoded by the signal processor 3704 with an externaldevice.

Further, the receiver 3700 includes an operation input unit 3710 thatreceives user operations as input. The receiver 3700 behaves inaccordance with control signals input by the operation input unit 3710according to user operations, such as by switching the power supply ONor OFF, changing the channel being received, switching subtitle displayON or OFF, switching between languages, changing the volume output bythe audio output unit 3706, and various other operations, includingmodifying the settings for receivable channels and the like.

The receiver 3700 may further include functionality for displaying anantenna level representing the received signal quality while thereceiver 3700 is receiving a signal. The antenna level may be, forexample, a index displaying the received signal quality calculatedaccording to the RSSI (Received Signal Strength Indicator), the receivedsignal magnetic field strength, the C/N (carrier-to-noise) ratio, theBER, the packet error rate, the frame error rate, the channel stateinformation, and so on, received by the receiver 3700 and indicating thelevel and the quality of a received signal. In such circumstances, thedemodulator 3702 includes a signal quality calibrator that measures theRSSI, the received signal magnetic field strength, the C/N ratio, theBER, the packet error rate, the frame error rate, the channel stateinformation, and so on. In response to user operations, the receiver3700 displays the antenna level (signal level, signal quality) in auser-recognizable format on the video display unit 3707. The displayformat for the antenna level (signal level, signal quality) may be anumerical value displayed according to the RSSI, the received signalmagnetic field strength, the C/N ratio, the BER, the packet error rate,the frame error rate, the channel state information, and so on, or maybe an image display that varies according to the RSSI, the receivedsignal magnetic field strength, the C/N ratio, the BER, the packet errorrate, the frame error rate, the channel state information, and so on.The receiver 3700 may display multiple antenna level (signal level,signal quality) calculated for each stream s1, s2, and so ondemultiplexed using the reception scheme discussed in theabove-described Embodiments, or may display a single antenna level(signal level, signal quality) calculated for all such streams. When thevideo data and audio data composing a program are transmittedhierarchically, the signal level (signal quality) may also be displayedfor each hierarchical level.

According to the above structure, the user is given an understanding ofthe antenna level (signal level, signal quality) numerically or visuallyduring reception using the reception schemes discussed in theabove-described Embodiments.

Although the above example describes the receiver 3700 as including theaudio output unit 3706, the video display unit 3707, the drive 3708, thestream output interface 3709, and the audiovisual output interface 3711,all of these components are not strictly necessary. As long as thereceiver 3700 includes at least one of the above-described components,the user is able to use the multiplexed data obtained throughdemodulation by the demodulator 3702 and error-correcting decoding. Anyreceiver may be freely combined with the above-described componentsaccording to the usage scheme.

(Multiplexed Data)

The following is a detailed description of a sample configuration ofmultiplexed data. The data configuration typically used in broadcastingis an MPEG-2 transport stream (TS). Therefore the following descriptiondescribes an example related to MPEG2-TS. However, the dataconfiguration of the multiplexed data transmitted by the transmissionand reception schemes discussed in the above-described Embodiments isnot limited to MPEG2-TS. The advantageous effects of the above-describedEmbodiments are also achievable using any other data structure.

FIG. 38 illustrates a sample configuration for multiplexed data. Asshown, the multiplexed data are elements making up programmes (orevents, being a portion thereof) currently provided by various services.For example, one or more video streams, audio streams, presentationgraphics (PG) streams, interactive graphics (IG) streams, and other suchelement streams are multiplexed to obtain the multiplexed data. When abroadcast program provided by the multiplexed data is a movie, the videostreams represent main video and sub video of the movie, the audiostreams represent main audio of the movie and sub-audio to be mixed withthe main audio, and the presentation graphics streams representsubtitles for the movie. Main video refers to video images normallypresented on a screen, whereas sub-video refers to video images (forexample, images of text explaining the outline of the movie) to bepresented in a small window inserted within the video images. Theinteractive graphics streams represent an interactive display made up ofGUI (Graphical User Interface) components presented on a screen.

Each stream included in the multiplexed data is identified by anidentifier, termed a PID, uniquely assigned to the stream. For example,PID 0x1011 is assigned to the video stream used for the main video ofthe movie, PIDs 0x1100 through 0x111F are assigned to the audio streams,PIDs 0x1200 through 0x121F are assigned to the presentation graphics,PIDs 0x1400 through 0x141F are assigned to the interactive graphics,PIDs 0x1B00 through 0x1B1F are assigned to the video streams used forthe sub-video of the movie, and PIDs 0x1A00 through 0x1A1F are assignedto the audio streams used as sub-audio to be mixed with the main audioof the movie.

FIG. 39 is a schematic diagram illustrating an example of themultiplexed data being multiplexed. First, a video stream 3901, made upof a plurality of frames, and an audio stream 3904, made up of aplurality of audio frames, are respectively converted into PES packetsequence 3902 and 3905, then further converted into TS packets 3903 and3906. Similarly, a presentation graphics stream 3911 and an interactivegraphics stream 3914 are respectively converted into PES packet sequence3912 and 3915, then further converted into TS packets 3913 and 3916. Themultiplexed data 3917 is made up of the TS packets 3903, 3906, 3913, and3916 multiplexed into a single stream.

FIG. 40 illustrates further details of a PES packet sequence ascontained in the video stream. The first tier of FIG. 40 shows a videoframe sequence in the video stream. The second tier shows a PES packetsequence. Arrows yy1, yy2, yy3, and yy4 indicate the plurality of VideoPresentation Units, which are I-pictures, B-pictures, and P-pictures, inthe video stream as divided and individually stored as the payload of aPES packet. Each PES packet has a PES header. A PES header contains aPTS (Presentation Time Stamp) at which the picture is to be displayed, aDTS (Decoding Time Stamp) at which the picture is to be decoded, and soon.

FIG. 41 illustrates the structure of a TS packet as ultimately writteninto the multiplexed data. A TS packet is a 188-byte fixed-length packetmade up of a 4-byte PID identifying the stream and of a 184-byte TSpayload containing the data. The above-described PES packets are dividedand individually stored as the TS payload. For a BD-ROM, each TS packethas a 4-byte TP_Extra_Header affixed thereto to build a 192-byte sourcepacket, which is to be written as the multiplexed data. TheTP_Extra_Header contains information such as an Arrival_Time_Stamp(ATS). The ATS indicates a time for starring transfer of the TS packetto the PID filter of a decoder. The multiplexed data are made up ofsource packets arranged as indicated in the bottom tier of FIG. 41. ASPN (source packet number) is incremented for each packet, beginning atthe head of the multiplexed data.

In addition to the video streams, audio streams, presentation graphicsstreams, and the like, the TS packets included in the multiplexed dataalso include a PAT (Program Association Table), a PMT (Program MapTable), a PCR (Program Clock Reference) and so on. The PAT indicates thePID of a PMT used in the multiplexed data, and the PID of the PAT itselfis registered as 0. The PMT includes PIDs identifying the respectivestreams, such as video, audio and subtitles, contained in themultiplexed data and attribute information (frame rate, aspect ratio,and the like) of the streams identified by the respective PIDs. Inaddition, the PMT includes various types of descriptors relating to themultiplexed data. One such descriptor may be copy control informationindicating whether or not copying of the multiplexed data is permitted.The PCR includes information for synchronizing the ATC (Arrival TimeClock) serving as the chronological axis of the ATS to the STC (SystemTime Clock) serving as the chronological axis of the PTS and DTS. EachPCR packet includes an STC time corresponding to the ATS at which thepacket is to be transferred to the decoder.

FIG. 42 illustrates the detailed data configuration of a PMT. The PMTstarts with a PMT header indicating the length of the data contained inthe PMT. Following the PMT header, descriptors pertaining to themultiplexed data are arranged. One example of a descriptor included inthe PMT is the copy control information described above. Following thedescriptors, stream information pertaining to the respective streamsincluded in the multiplexed data is arranged. Each piece of streaminformation is composed of stream descriptors indicating a stream typeidentifying a compression codec employed for a corresponding stream, aPID for the stream, and attribute information (frame rate, aspect ratio,and the like) of the stream. The PMT includes the same number of streamdescriptors as the number of streams included in the multiplexed data.

When recorded onto a recoding medium or the like, the multiplexed dataare recorded along with a multiplexed data information file.

FIG. 43 illustrates a sample configuration for the multiplexed datainformation file. As shown, the multiplexed data information file ismanagement information for the multiplexed data, is provided inone-to-one correspondence with the multiplexed data, and is made up ofmultiplexed data information, stream attribute information, and an entrymap.

The multiplexed data information is made up of a system rate, a playbackstart time, and a playback end time. The system rate indicates themaximum transfer rate of the multiplexed data to the PID filter of alater-described system target decoder. The multiplexed data includes ATSat an interval set so as not to exceed the system rate. The playbackstart time is set to the time specified by the PTS of the first videoframe in the multiplexed data, whereas the playback end time is set tothe time calculated by adding the playback duration of one frame to thePTS of the last video frame in the multiplexed data.

FIG. 44 illustrates a sample configuration for the stream attributeinformation included in the multiplexed data information file. As shown,the stream attribute information is attribute information for eachstream included in the multiplexed data, registered for each PID. Thatis, different pieces of attribute information are provided for differentstreams, namely for the video streams, the audio streams, thepresentation graphics streams, and the interactive graphics streams. Thevideo stream attribute information indicates the compression codecemployed to compress the video stream, the resolution of individualpictures constituting the video stream, the aspect ratio, the framerate, and so on. The audio stream attribute information indicates thecompression codec employed to compress the audio stream, the number ofchannels included in the audio stream, the language of the audio stream,the sampling frequency, and so on. This information is used toinitialize the decoder before playback by a player.

In the present Embodiment, the stream type included in the PMT is usedamong the information included in the multiplexed data. When themultiplexed data are recorded on a recording medium, the video streamattribute information included in the multiplexed data information fileis used. Specifically, the video coding method and device described inany of the above Embodiments may be modified to additionally include astep or unit of setting a specific piece of information in the streamtype included in the PMT or in the video stream attribute information.The specific piece of information is for indicating that the video dataare generated by the video coding method and device described in theEmbodiment. According to such a structure, video data generated by thevideo coding method and device described in any of the above Embodimentsis distinguishable from video data compliant with other standards.

FIG. 45 illustrates a sample configuration of an audiovisual outputdevice 4500 that includes a reception device 4504 receiving a modulatedsignal that includes audio and video data transmitted by a broadcaster(base station) or data intended for broadcasting. The configuration ofthe reception device 4504 corresponds to the reception device 3700 fromFIG. 37. The audiovisual output device 4500 incorporates, for example,an OS (Operating System), or incorporates a communication device 4506for connecting to the Internet (e.g., a communication device intendedfor a wireless LAN (Local Area Network) or for Ethernet™). As such, avideo display unit 4501 is able to simultaneously display audio andvideo data, or video in video data for broadcast 4502, and hypertext4503 (from the World Wide Web) provided over the Internet. By operatinga remote control 4507 (alternatively, a mobile phone or keyboard),either of the video in video data for broadcast 4502 and the hypertext4503 provided over the Internet may be selected to change operations.For example, when the hypertext 4503 provided over the Internet isselected, the website displayed may be changed by remote controloperations. When audio and video data, or video in video data forbroadcast 4502 is selected, information from a selected channel(selected (television) program or audio broadcast) may be transmitted bythe remote control 4507. As such, an interface 4505 obtains theinformation transmitted by the remote control. The reception device 4504performs processing such as demodulation and error-correctioncorresponding to the selected channel, thereby obtaining the receiveddata. At this point, the reception device 4504 obtains control symbolinformation that includes information on the transmission scheme (asdescribed using FIG. 5) from control symbols included the signalcorresponding to the selected channel. As such, the reception device4504 is able to correctly set the reception operations, demodulationscheme, error-correction scheme and so on, thus enabling the dataincluded in the data symbols transmitted by the broadcaster (basestation) to be obtained. Although the above description is given for anexample of the user using the remote control 4507, the same operationsapply when the user presses a selection key embedded in the audiovisualoutput device 4500 to select a channel.

In addition, the audiovisual output device 4500 may be operated usingthe Internet. For example, the audiovisual output device 4500 may bemade to record (store) a program through another terminal connected tothe Internet. (Accordingly, the audiovisual output device 4500 shouldinclude the drive 3708 from FIG. 37.) The channel is selected beforerecording begins. As such, the reception device 4504 performs processingsuch as demodulation and error-correction corresponding to the selectedchannel, thereby obtaining the received data. At this point, thereception device 4504 obtains control symbol information that includesinformation on the transmission scheme (the transmission scheme,modulation scheme, error-correction scheme, and so on from theabove-described Embodiments) (as described using FIG. 5) from controlsymbols included the signal corresponding to the selected channel. Assuch, the reception device 4504 is able to correctly set the receptionoperations, demodulation scheme, error-correction scheme and so on, thusenabling the data included in the data symbols transmitted by thebroadcaster (base station) to be obtained.

(Supplement)

The present description considers a communications/broadcasting devicesuch as a broadcaster, a base station, an access point, a terminal, amobile phone, or the like provided with the transmission device, and acommunications device such as a television, radio, terminal, personalcomputer, mobile phone, access point, base station, or the like providedwith the reception device. The transmission device and the receptiondevice pertaining to the present invention are communication devices ina form able to execute applications, such as a television, radio,personal computer, mobile phone, or similar, through connection to somesort of interface (e.g., USB).

Furthermore, in the present Embodiment, symbols other than data symbols,such as pilot symbols (namely preamble, unique word, postamble,reference symbols, scattered pilot symbols and so on), symbols intendedfor control information, and so on may be freely arranged within theframe. Although pilot symbols and symbols intended for controlinformation are presently named, such symbols may be freely namedotherwise as the function thereof remains the important consideration.

Provided that a pilot symbol, for example, is a known symbol modulatedwith PSK modulation in the transmitter and receiver (alternatively, thereceiver may be synchronized such that the receiver knows the symbolstransmitted by the transmitter), the receiver is able to use this symbolfor frequency synchronization, time synchronization, channel estimation(CSI (Channel State Information) estimation for each modulated signal),signal detection, and the like.

The symbols intended for control information are symbols transmittinginformation (such as the modulation scheme, error-correcting codingscheme, coding rate of error-correcting codes, and setting informationfor the top layer used in communications) transmitted to the receivingparty in order to execute transmission of non-data (i.e., applications).

The present invention is not limited to the Embodiments, but may also berealized in various other ways. For example, while the above Embodimentsdescribe communication devices, the present invention is not limited tosuch devices and may be implemented as software for the correspondingcommunications scheme.

Although the above-described Embodiments describe phase changing schemesfor schemes of transmitting two modulated signals from two antennas, nolimitation is intended in this regard. Precoding and a change of phasemay be performed on four signals that have been mapped to generate fourmodulated signals transmitted using four antennas. That is, the presentinvention is applicable to performing a change of phase on N signalsthat have been mapped and precoded to generate N modulated signalstransmitted using N antennas.

Although the above-described Embodiments describe examples of systemswhere two modulated signals are transmitted from two antennas andreceived by two respective antennas in a MIMO system, the presentinvention is not limited in this regard and is also applicable to MISO(Multiple Input Single Output) systems. In a MISO system, the receptiondevice does not include antenna 701_Y, wireless unit 703_Y, channelfluctuation estimator 707_1 for modulated signal z1, and channelfluctuation estimator 707_2 for modulated signal z2 from FIG. 7.However, the processing described in Embodiment 1 may still be executedto estimate r1 and r2. Technology for receiving and decoding a pluralityof signals transmitted simultaneously at a common frequency are receivedby a single antenna is widely known. The present invention is additionalprocessing supplementing conventional technology for a signal processorreverting a phase changed by the transmitter.

Although the present invention describes examples of systems where twomodulated signals are transmitted from two antennas and received by tworespective antennas in a MIMO communications system, the presentinvention is not limited in this regard and is also applicable to MISOsystems. In a MISO system, the transmission device performs precodingand change of phase such that the points described thus far areapplicable. However, the reception device does not include antenna701_Y, wireless unit 703_Y, channel fluctuation estimator 707_1 formodulated signal z1, and channel fluctuation estimator 707_2 formodulated signal z2 from FIG. 7. However, the processing described inthe present description may still be executed to estimate the datatransmitted by the transmission device. Technology for receiving anddecoding a plurality of signals transmitted simultaneously at a commonfrequency are received by a single antenna is widely known (asingle-antenna receiver may apply ML operations (Max-log APP orsimilar)). The present invention may have the signal processor 711 fromFIG. 7 perform demodulation (detection) by taking the precoding andchange of phase applied by the transmitter into consideration.

The present description uses terms such as precoding, precoding weights,precoding matrix, and so on. The terminology itself may be otherwise(e.g., may be alternatively termed a codebook) as the key point of thepresent invention is the signal processing itself.

Furthermore, although the present description discusses examples mainlyusing OFDM as the transmission scheme, the invention is not limited inthis manner. Multi-carrier schemes other than OFDM and single-carrierschemes may all be used to achieve similar Embodiments. Here,spread-spectrum communications may also be used. When single-carrierschemes are used, a change of phase is performed with respect to thetime domain.

In addition, although the present description discusses the use of MLoperations, APP, Max-log APP, ZF, MMSE and so on by the receptiondevice, these operations may all be generalized as wave detection,demodulation, detection, estimation, and demultiplexing as the softresults (log-likelihood and log-likelihood ratio) and the hard results(zeroes and ones) obtained thereby are the individual bits of datatransmitted by the transmission device.

Different data may be transmitted by each stream s1(t) and s2(t) (s1(i),s2(i)), or identical data may be transmitted thereby.

The two stream baseband signals s1(i) and s2(i) (where i indicatessequence (with respect to time or (carrier) frequency)) undergoprecoding and a regular change of phase (the order of operations may befreely reversed) to generate two post-processing baseband signals z1(i)and z2(i). For post-processing baseband signal z1(i), the in-phasecomponent I is I₁(i) while the quadrature component is Q₁(i), and forpost processing baseband signal z2(i), the in-phase component is I₁(i)while the quadrature component is Q₂(i). The baseband components may beswitched, as long as the following holds.

Let the in-phase component and the quadrature component of switchedbaseband signal r1(i) be I₁(i) and Q₂(i), and the in-phase component andthe quadrature component of switched baseband signal r2(i) be I₂(i) andQ₁(i). The modulated signal corresponding to switched baseband signalr1(i) is transmitted by transmit antenna 1 and the modulated signalcorresponding to switched baseband signal r2(i) is transmitted fromtransmit antenna 2, simultaneously on a common frequency. As such, themodulated signal corresponding to switched baseband signal r1(i) and themodulated signal corresponding to switched baseband signal r2(i) aretransmitted from different antennas, simultaneously on a commonfrequency. Alternatively,

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r1(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r1(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r1(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r2(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r2(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be Q₂(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be I₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r1(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be I₁(i)while the quadrature component may be Q₂(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i).

For switched baseband signal r2(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be I₂(i) while the quadraturecomponent may be Q₁(i).

For switched baseband signal r2(i), the in-phase component may be Q₂(i)while the quadrature component may be I₁(i), and for switched basebandsignal r1(i), the in-phase component may be Q₁(i) while the quadraturecomponent may be I₂(i). Alternatively, although the above descriptiondiscusses performing two types of signal processing on both streamsignals so as to switch the in-phase component and quadrature componentof the two signals, the invention is not limited in this manner. The twotypes of signal processing may be performed on more than two streams, soas to switch the in-phase component and quadrature component thereof.

Alternatively, although the above examples describe switching basebandsignals having a common time (common (sub-)carrier) frequency), thebaseband signals being switched need not necessarily have a common time.For example, any of the following are possible.

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r1(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r1(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r1(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r2(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be Q₂(i+w).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be I₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beI₁(i+v) while the quadrature component may be Q₂(i+w), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

For switched baseband signal r2(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be I₂(i+w) while thequadrature component may be Q₁(i+v).

For switched baseband signal r2(i), the in-phase component may beQ₂(i+w) while the quadrature component may be I₁(i+v), and for switchedbaseband signal r1(i), the in-phase component may be Q₁(i+v) while thequadrature component may be I₂(i+w).

FIG. 55 illustrates a baseband signal switcher 5502 explaining theabove. As shown, of the two processed baseband signals z1(i) 5501_1 andz2(i) 5501_2, processed baseband signal z1(i) 5501_1 has in-phasecomponent I₁(i) and quadrature component Q₁(i), while processed basebandsignal z2(i) 55012 has in-phase component I₂(i) and quadrature componentQ₂(i). Then, after switching, switched baseband signal r1(i) 5503_1 hasin-phase component I_(r1)(i) and quadrature component Q_(r1)(i), whileswitched baseband signal r2(i) 5503_2 has in-phase component I_(r2)(i)and quadrature component Q_(r2)(i). The in-phase component I_(r1)(i) andquadrature component Q_(r)i(i) of switched baseband signal r1(i) 5503_1and the in-phase component Ir2(i) and quadrature component Q_(r2)(i) ofswitched baseband signal r2(i) 5503_2 may be expressed as any of theabove. Although this example describes switching performed on basebandsignals having a common time (common ((sub-)carrier) frequency) andhaving undergone two types of signal processing, the same may be appliedto baseband signals having undergone two types of signal processing buthaving different time (different ((sub-)carrier) frequencies).

Each of the transmit antennas of the transmission device and each of thereceive antennas of the reception device shown in the figures may beformed by a plurality of antennas.

The present description uses the symbol ∀, which is the universalquantifier, and the symbol ∃, which is the existential quantifier.

Furthermore, the present description uses the radian as the unit ofphase in the complex plane, e.g., for the argument thereof.

When dealing with the complex plane, the coordinates of complex numbersare expressible by way of polar coordinates. For a complex numberz=a+j_(b) (where a and b are real numbers and j is the imaginary unit),the corresponding point (a, b) on the complex plane is expressed withthe polar coordinates[r, θ], converted as follows:a=r×cos θb=r×sin θ[Math. 49]r=√{square root over (a ² +b ²)}  (formula 49)

where r is the absolute value of z (r=|z|), and θ is the argumentthereof. As such, z=a+j_(b) is expressible as re^(jθ).

In the present invention, the baseband signals s1, s2, z1, and z2 aredescribed as being complex signals. A complex signal made up of in-phasesignal I and quadrature signal Q is also expressible as complex signalI+jQ. Here, either of I and Q may be equal to zero.

FIG. 46 illustrates a sample broadcasting system using the phasechanging scheme described in the present description. As shown, a videoencoder 4601 takes video as input, performs video encoding, and outputsencoded video data 4602. An audio encoder takes audio as input, performsaudio encoding, and outputs encoded audio data 4604. A data encoder 4605takes data as input, performs data encoding (e.g., data compression),and outputs encoded data 4606. Taken as a whole, these components form asource information encoder 4600.

A transmitter 4607 takes the encoded video data 4602, the encoded audiodata 4604, and the encoded data 4606 as input, performs error-correctingcoding, modulation, precoding, and phase changing (e.g., the signalprocessing by the transmission device from FIG. 3) on a subset of or onthe entirety of these, and outputs transmit signals 4608_1 through4608_N. Transmit signals 4608_1 through 4608_N are then transmitted byantennas 4609_1 through 4609_N as radio waves.

A receiver 4612 takes received signals 4611_1 through 4611_M received byantennas 4610_1 through 4610_M as input, performs processing such asfrequency conversion, change of phase, decoding of the precoding,log-likelihood ratio calculation, and error-correcting decoding (e.g.,the processing by the reception device from FIG. 7), and outputsreceived data 4613, 4615, and 4617. A source information decoder 4619takes the received data 4613, 4615, and 4617 as input. A video decoder4614 takes received data 4613 as input, performs video decoding, andoutputs a video signal. The video is then displayed on a televisiondisplay. An audio decoder 4616 takes received data 4615 as input. Theaudio decoder 4616 performs audio decoding and outputs an audio signal.the audio is then played through speakers. A data decoder 4618 takesreceived data 4617 as input, performs data decoding, and outputsinformation.

In the above-described Embodiments pertaining to the present invention,the number of encoders in the transmission device using a multi-carriertransmission scheme such as OFDM may be any number, as described above.Therefore, as in FIG. 4, for example, the transmission device may haveonly one encoder and apply a scheme for distributing output to themulti-carrier transmission scheme such as OFDM. In such circumstances,the wireless units 310A and 310B from FIG. 4 should replace theOFDM-related processors 1301A and 1301B from FIG. 12. The description ofthe OFDM-related processors is as given for Embodiment 1.

Although Embodiment 1 gives formula 36 as an example of a precodingmatrix, another precoding matrix may also be used, when the followingscheme is applied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 50} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\;\pi}} \\{\alpha \times e^{j\; 0}} & e^{j\; 0}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 50} \right)\end{matrix}$

In the precoding matrices of formula 36 and formula 50, the value of αis set as given by formula 37 and formula 38. However, no limitation isintended in this manner. A simple precoding matrix is obtainable bysetting α=1, which is also a valid value.

In Embodiment A1, the phase changers from FIGS. 3, 4, 6, 12, 25, 29, 51,and 53 are indicated as having a phase changing value of PHASE[i] (wherei=0, 1, 2 . . . N−2, N−1 (i denotes an integer that satisfies 0≤i≤N−1))to achieve a period (cycle) of N (value reached given that FIGS. 3, 4,6, 12, 25, 29, 51, and 53 perform a change of phase on only one basebandsignal). The present description discusses performing a change of phaseon one precoded baseband signal (i.e., in FIGS. 3, 4, 6, 12, 25, 29, and51) namely on precoded baseband signal z2′. Here, PHASE[k] is calculatedas follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 51} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {\frac{2\; k\;\pi}{N}\mspace{14mu}{radians}}} & \left( {{formula}\mspace{14mu} 51} \right)\end{matrix}$

where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≤k≤N−1). When N=5, 7, 9, 11, or 15, the reception device is able toobtain good data reception quality.

Although the present description discusses the details of phase changingschemes involving two modulated signals transmitted by a plurality ofantennas, no limitation is intended in this regard. Precoding and achange of phase may be performed on three or more baseband signals onwhich mapping has been performed according to a modulation scheme,followed by predetermined processing on the post-phase-change basebandsignals and transmission using a plurality of antennas, to realize thesame results.

Programs for executing the above transmission scheme may, for example,be stored in advance in ROM (Read-Only Memory) and be read out foroperation by a CPU.

Furthermore, the programs for executing the above transmission schememay be stored on a computer-readable recording medium, the programsstored in the recording medium may be loaded in the RAM (Random AccessMemory) of the computer, and the computer may be operated in accordancewith the programs.

The components of the above-described Embodiments may be typicallyassembled as an LSI (Large Scale Integration), a type of integratedcircuit. Individual components may respectively be made into discretechips, or a subset or entirety of the components may be made into asingle chip. Although an LSI is mentioned above, the terms IC(Integrated Circuit), system LSI, super LSI, or ultra LSI may alsoapply, depending on the degree of integration. Furthermore, the methodof integrated circuit assembly is not limited to LSI. A dedicatedcircuit or a general-purpose processor may be used. After LSI assembly,a FPGA (Field Programmable Gate Array) or reconfigurable processor maybe used.

Furthermore, should progress in the field of semiconductors or emergingtechnologies lead to replacement of LSI with other integrated circuitmethods, then such technology may of course be used to integrate thefunctional blocks. Applications to biotechnology are also plausible.

Embodiment C1

Embodiment 1 explained that the precoding matrix in use may be switchedwhen transmission parameters change. The present Embodiment describes adetailed example of such a case, where, as described above (in thesupplement), the transmission parameters change such that streams s1(t)and s2(t) switch between transmitting different data and transmittingidentical data, and the precoding matrix and phase changing scheme beingused are switched accordingly.

The example of the present Embodiment describes a situation where twomodulated signals transmitted from two different transmit antennaalternate between having the modulated signals include identical dataand having the modulated signals each include different data.

FIG. 56 illustrates a sample configuration of a transmission deviceswitching between transmission schemes, as described above. In FIG. 56,components operating in the manner described for FIG. 54 use identicalreference numbers. As shown, FIG. 56 differs from FIG. 54 in that adistributor 404 takes the frame configuration signal 313 as input. Theoperations of the distributor 404 are described using FIG. 57.

FIG. 57 illustrates the operations of the distributor 404 whentransmitting identical data and when transmitting different data. Asshown, given encoded data x1, x2, x3, x4, x5, x6, and so on, whentransmitting identical data, distributed data 405 is given as x1, x2,x3, x4, x5, x6, and so on, while distributed data 405B is similarlygiven as x1, x2, x3, x4, x5, x6, and so on.

On the other hand, when transmitting different data, distributed data405A are given as x1, x3, x5, x7, x9, and so on, while distributed data405B are given as x2, x4, x6, x8, x10, and so on.

The distributor 404 determines, according to the frame configurationsignal 313 taken as input, whether the transmission mode is identicaldata transmission or different data transmission.

An alternative to the above is shown in FIG. 58. As shown, whentransmitting identical data, the distributor 404 outputs distributeddata 405A as x1, x2, x3, x4, x5, x6, and so on, while outputting nothingas distributed data 405B. Accordingly, when the frame configurationsignal 313 indicates identical data transmission, the distributor 404operates as described above, while interleaver 304B and mapper 306B fromFIG. 56 do not operate. Thus, only baseband signal 307A output by mapper306A from FIG. 56 is valid, and is taken as input by both weighting unit308A and 308B.

One characteristic feature of the present Embodiment is that, when thetransmission mode switches from identical data transmission to differentdata transmission, the precoding matrix may also be switched. Asindicated by formula 36 and formula 39 in Embodiment 1, given a matrixmade up of w11, w12, w21, and w22, the precoding matrix used to transmitidentical data may be as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 52} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = \begin{pmatrix}a & 0 \\0 & a\end{pmatrix}} & \left( {{formula}\mspace{14mu} 52} \right)\end{matrix}$

where a is a real number (a may also be a complex number, but given thatthe baseband signal input as a result of precoding undergoes a change ofphase, a real number is preferable for considerations of circuit sizeand complexity reduction). Also, when a is equal to one, the weightingunits 308A and 308B do not perform weighting and output the input signalas-is.

Accordingly, when transmitting identical data, the weighted basebandsignals 309A and 316B are identical signals output by the weightingunits 308A and 308B.

When the frame configuration signal indicates identical transmissionmode, a phase changer 5201 performs a change of phase on weightedbaseband signal 309A and outputs post-phase-change baseband signal 5202.Similarly, when the frame configuration signal indicates identicaltransmission mode, phase changer 317B performs a change of phase onweighted baseband signal 316B and outputs post-phase-change basebandsignal 309B. The change of phase performed by phase changer 5201 is ofe^(jA(t)) (alternatively, e^(jA(f)) or e^(jA(t,f))) (where t is time andf is frequency) (accordingly, e^(jA(t)) (alternatively, e^(jA(f)) ore^(jA(t,f))) is the value by which the input baseband signal ismultiplied), and the change of phase performed by phase changer 317B isof ejB(t) (alternatively, e^(jB(f)) or e^(jB(t,f))) (where t is time andf is frequency) (accordingly, e^(jB(t)) (alternatively, e^(jB(f)) ore^(jB(t,f))) is the value by which the input baseband signal ismultiplied). As such, the following condition is satisfied.[Math. 53]

Some time t satisfiese ^(jA(t)) ≠e ^(jB(t))  (formula 53)

(Or, some (carrier) frequency f satisfies e^(jA(f))≠e^(jB(f)))

(Or, some (carrier) frequency f and time t satisfye^(jA(t,f))≠e^(jB(t,f)))

As such, the transmit signal is able to reduce multi-path influence andthereby improve data reception quality for the reception device.(However, the change of phase may also be performed by only one of theweighted baseband signals 309A and 316B.)

In FIG. 56, when OFDM is used, processing such as IFFT and frequencyconversion is performed on post-phase-change baseband signal 5202, andthe result is transmitted by a transmit antenna. (See FIG. 13)(Accordingly, post-phase-change baseband signal 5202 may be consideredthe same as signal 1301A from FIG. 13.) Similarly, when OFDM is used,processing such as IFFT and frequency conversion is performed onpost-phase-change baseband signal 309B, and the result is transmitted bya transmit antenna. (See FIG. 13) (Accordingly, post-phase-changebaseband signal 309B may be considered the same as signal 1301B fromFIG. 13.)

When the selected transmission mode indicates different datatransmission, then any of formula 36, formula 39, and formula 50 givenin Embodiment 1 may apply. Significantly, the phase changers 5201 and317B from FIG. 56 us a different phase changing scheme than whentransmitting identical data. Specifically, as described in Embodiment 1,for example, phase changer 5201 performs the change of phase while phasechanger 317B does not, or phase changer 317B performs the change ofphase while phase changer 5201 does not. Only one of the two phasechangers performs the change of phase. As such, the reception deviceobtains good data reception quality in the LOS environment as well asthe NLOS environment.

When the selected transmission mode indicates different datatransmission, the precoding matrix may be as given in formula 52, or asgiven in any of formula 36, formula 50, and formula 39, or may be aprecoding matrix unlike that given in formula 52. Thus, the receptiondevice is especially likely to experience improvements to data receptionquality in the LOS environment.

Furthermore, although the present Embodiment discusses examples usingOFDM as the transmission scheme, the invention is not limited in thismanner. Multi-carrier schemes other than OFDM and single-carrier schemesmay all be used to achieve similar Embodiments. Here, spread-spectrumcommunications may also be used. When single-carrier schemes are used,the change of phase is performed with respect to the time domain.

As explained in Embodiment 3, when the transmission scheme involvesdifferent data transmission, the change of phase is performed on thedata symbols, only. However, as described in the present Embodiment,when the transmission scheme involves identical data transmission, thenthe change of phase need not be limited to the data symbols but may alsobe performed on pilot symbols, control symbols, and other such symbolsinserted into the transmission frame of the transmit signal. (The changeof phase need not always be performed on symbols such as pilot symbolsand control symbols, though doing so is preferable in order to achievediversity gain.)

Embodiment C2

The present Embodiment describes a configuration scheme for a basestation corresponding to Embodiment C1.

FIG. 59 illustrates the relationship of a base stations (broadcasters)to terminals. A terminal P (5907) receives transmit signal 5903Atransmitted by antenna 5904A and transmit signal 5905A transmitted byantenna 5906A of broadcaster A (5902A), then performs predeterminedprocessing thereon to obtained received data.

A terminal Q (5908) receives transmit signal 5903A transmitted byantenna 5904A of base station A (5902A) and transmit signal 593Btransmitted by antenna 5904B of base station B (5902B), then performspredetermined processing thereon to obtained received data.

FIGS. 60 and 61 illustrate the frequency allocation of base station A(5902A) for transmit signals 5903A and 5905A transmitted by antennas5904A and 5906A, and the frequency allocation of base station B (5902B)for transmit signals 5903B and 5905B transmitted by antennas 5904B and5906B. In FIGS. 60 and 61, frequency is on the horizontal axis andtransmission power is on the vertical axis.

As shown, transmit signals 5903A and 5905A transmitted by base station A(5902A) and transmit signals 5903B and 5905B transmitted by base stationB (5902B) use at least frequency band X and frequency band Y. Frequencyband X is used to transmit data of a first channel, and frequency band Yis used to transmit data of a second channel.

Accordingly, terminal P (5907) receives transmit signal 5903Atransmitted by antenna 5904A and transmit signal 5905A transmitted byantenna 5906A of base station A (5902A), extracts frequency band Xtherefrom, performs predetermined processing, and thus obtains the dataof the first channel. Terminal Q (5908) receives transmit signal 5903Atransmitted by antenna 5904A of base station A (5902A) and transmitsignal 5903B transmitted by antenna 5904B of base station B (5902B),extracts frequency band Y therefrom, performs predetermined processing,and thus obtains the data of the second channel.

The following describes the configuration and operations of base stationA (5902A) and base station B (5902B).

As described in Embodiment C1, both base station A (5902A) and basestation B (5902B) incorporate a transmission device configured asillustrated by FIGS. 56 and 13. When transmitting as illustrated by FIG.60, base station A (5902A) generates two different modulated signals (onwhich precoding and a change of phase are performed) with respect tofrequency band X as described in Embodiment C1. The two modulatedsignals are respectively transmitted by the antennas 5904A and 5906A.With respect to frequency band Y, base station A (5902A) operatesinterleaver 304A, mapper 306A, weighting unit 308A, and phase changerfrom FIG. 56 to generate modulated signal 5202. Then, a transmit signalcorresponding to modulated signal 5202 is transmitted by antenna 1310Afrom FIG. 13, i.e., by antenna 5904A from FIG. 59. Similarly, basestation B (5902B) operates interleaver 304A, mapper 306A, weighting unit308A, and phase changer 5201 from FIG. 56 to generate modulated signal5202. Then, a transmit signal corresponding to modulated signal 5202 istransmitted by antenna 1310A from FIG. 13, i.e., by antenna 5904B fromFIG. 59.

The creation of encoded data in frequency band Y may involve, as shownin FIG. 56, generating encoded data in individual base stations or mayinvolve having one of the base stations generate such encoded data fortransmission to other base stations. As an alternative scheme, one ofthe base stations may generate modulated signals and be configured topass the modulated signals so generated to other base stations.

Also, in FIG. 59, signal 5901 includes information pertaining to thetransmission mode (identical data transmission or different datatransmission). The base stations obtain this signal and thereby switchbetween generation schemes for the modulated signals in each frequencyband. Here, signal 5901 is indicated in FIG. 59 as being input fromanother device or from a network. However, configurations where, forexample, base station A (5902) is a master station passing a signalcorresponding to signal 5901 to base station B (5902B) are alsopossible.

As explained above, when the base station transmits different data, theprecoding matrix and phase changing scheme are set according to thetransmission scheme to generate modulated signals.

On the other hand, to transmit identical data, two base stationsrespectively generate and transmit modulated signals. In suchcircumstances, base stations each generating modulated signals fortransmission from a common antenna may be considered to be two combinedbase stations using the precoding matrix given by formula 52. The phasechanging scheme is as explained in Embodiment C1, for example, andsatisfies the conditions of formula 53.

In addition, the transmission scheme of frequency band X and frequencyband Y may vary over time. Accordingly, as illustrated in FIG. 61, astime passes, the frequency allocation changes from that indicated inFIG. 60 to that indicated in FIG. 61.

According to the present Embodiment, not only can the reception deviceobtain improved data reception quality for identical data transmissionas well as different data transmission, but the transmission devices canalso share a phase changer.

Furthermore, although the present Embodiment discusses examples usingOFDM as the transmission scheme, the invention is not limited in thismanner. Multi-carrier schemes other than OFDM and single-carrier schemesmay all be used to achieve similar Embodiments. Here, spread-spectrumcommunications may also be use. When single-carrier schemes are used,the change of phase is performed with respect to the time domain.

As explained in Embodiment 3, when the transmission scheme involvesdifferent data transmission, the change of phase is carried out on thedata symbols, only. However, as described in the present Embodiment,when the transmission scheme involves identical data transmission, thenthe change of phase need not be limited to the data symbols but may alsobe performed on pilot symbols, control symbols, and other such symbolsinserted into the transmission frame of the transmit signal. (The changeof phase need not always be performed on symbols such as pilot symbolsand control symbols, though doing so is preferable in order to achievediversity gain.)

Embodiment C3

The present Embodiment describes a configuration scheme for a repeatercorresponding to Embodiment C1. The repeater may also be termed arepeating station.

FIG. 62 illustrates the relationship of a base stations (broadcasters)to repeaters and terminals. As shown in FIG. 63, base station 6201 atleast transmits modulated signals on frequency band X and frequency bandY. Base station 6201 transmits respective modulated signals on antenna6202A and antenna 6202B. The transmission scheme here used is describedlater, with reference to FIG. 63.

Repeater A (6203A) performs processing such as demodulation on receivedsignal 6205A received by receive antenna 6204A and on received signal6207A received by receive antenna 6206A, thus obtaining received data.Then, in order to transmit the received data to a terminal, repeater A(6203A) performs transmission processing to generate modulated signals6209A and 6211A for transmission on respective antennas 6210A and 6212A.

Similarly, repeater B (6203B) performs processing such as demodulationon received signal 6205B received by receive antenna 6204B and onreceived signal 6207B received by receive antenna 6206B, thus obtainingreceived data. Then, in order to transmit the received data to aterminal, repeater B (6203B) performs transmission processing togenerate modulated signals 6209B and 6211B for transmission onrespective antennas 6210B and 6212B. Here, repeater B (6203B) is amaster repeater that outputs a control signal 6208. repeater A (6203A)takes the control signal as input. A master repeater is not strictlynecessary. Base station 6201 may also transmit individual controlsignals to repeater A (6203A) and to repeater B (6203B).

Terminal P (5907) receives modulated signals transmitted by repeater A(6203A), thereby obtaining data. Terminal Q (5908) receives signalstransmitted by repeater A (6203A) and by repeater B (6203B), therebyobtaining data. Terminal R (6213) receives modulated signals transmittedby repeater B (6203B), thereby obtaining data.

FIG. 63 illustrates the frequency allocation for a modulated signaltransmitted by antenna 6202A among transmit signals transmitted by thebase station, and the frequency allocation of modulated signalstransmitted by antenna 6202B. In FIG. 63, frequency is on the horizontalaxis and transmission power is on the vertical axis.

As shown, the modulated signals transmitted by antenna 6202A and byantenna 6202B use at least frequency band X and frequency band Y.Frequency band X is used to transmit data of a first channel, andfrequency band Y is used to transmit data of a second channel.

As described in Embodiment C1, the data of the first channel istransmitted using frequency band X in different data transmission mode.Accordingly, as shown in FIG. 63, the modulated signals transmitted byantenna 6202A and by antenna 6202B include components of frequency bandX. These components of frequency band X are received by repeater A andby repeater B. Accordingly, as described in Embodiment 1 and inEmbodiment C1, modulated signals in frequency band X are signals onwhich mapping has been performed, and to which precoding (weighting) andthe change of phase are applied.

As shown in FIG. 62, the data of the second channel is transmitted byantenna 6202A of FIG. 2 and transmits data in components of frequencyband Y. These components of frequency band Y are received by repeater Aand by repeater B.

FIG. 64 illustrate the frequency allocation for transmit signalstransmitted by repeater A and repeater B, specifically for modulatedsignal 6209A transmitted by antenna 6210A and modulated signal 6211Atransmitted by antenna 6212A of repeater 6210A, and for modulated signal6209B transmitted by antenna 6210B and modulated signal 6211Btransmitted by antenna 6212B of repeater B. In FIG. 64, frequency is onthe horizontal axis and transmission power is on the vertical axis.

As shown, modulated signal 6209A transmitted by antenna 6210A andmodulated signal 6211A transmitted by antenna 6212A use at leastfrequency band X and frequency band Y. Also, modulated signal 6209Btransmitted by antenna 6210B and modulated signal 6211B transmitted byantenna 6212B similarly use at least frequency band X and frequency bandY. Frequency band X is used to transmit data of a first channel, andfrequency band Y is used to transmit data of a second channel.

As described in Embodiment C1, the data of the first channel istransmitted using frequency band X in different data transmission mode.Accordingly, as shown in FIG. 64, modulated signal 6209A transmitted byantenna 6210A and modulated signal 6211A transmitted by antenna 6212Binclude components of frequency band X. These components of frequencyband X are received by terminal P. Similarly, as shown in FIG. 64,modulated signal 6209B transmitted by antenna 6210B and modulated signal6211B transmitted by antenna 6212B include components of frequency bandX. These components of frequency band X are received by terminal R.Accordingly, as described in Embodiment 1 and in Embodiment C1,modulated signals in frequency band X are signals on which mapping hasbeen performed, and to which precoding (weighting) and the change ofphase are applied.

As shown in FIG. 64, the data of the second channel is carried by themodulated signals transmitted by antenna 6210A of repeater A (6203A) andby antenna 6210B of repeater B (6203) from FIG. 62 and transmits data incomponents of frequency band Y. Here, the components of frequency band Yin modulated signal 6209A transmitted by antenna 6210A of repeater A(6203A) and those in modulated signal 6209B transmitted by antenna 6210Bof repeater B (6203B) are used in a transmission mode that involvesidentical data transmission, as explained in Embodiment C1. Thesecomponents of frequency band Y are received by terminal Q.

The following describes the configuration of repeater A (6203A) andrepeater B (6203B) from FIG. 62, with reference to FIG. 65.

FIG. 65 illustrates a sample configuration of a receiver and transmitterin a repeater. Components operating identically to those of FIG. 56 usethe same reference numbers thereas. Receiver 6203X takes received signal6502A received by receive antenna 6501A and received signal 6502Breceived by receive antenna 6501B as input, performs signal processing(signal demultiplexing or compositing, error-correction decoding, and soon) on the components of frequency band X thereof to obtain data 6204Xtransmitted by the base station using frequency band X, outputs the datato the distributor 404 and obtains transmission scheme informationincluded in control information (and transmission scheme informationwhen transmitted by a repeater), and outputs the frame configurationsignal 313.

Receiver 6203X and onward constitute a processor for generating amodulated signal for transmitting frequency band X. Further, thereceiver here described is not only the receiver for frequency band X asshown in FIG. 65, but also incorporates receivers for other frequencybands. Each receiver forms a processor for generating modulated signalsfor transmitting a respective frequency band.

The overall operations of the distributor 404 are identical to those ofthe distributor in the base station described in Embodiment C2.

When transmitting as indicated in FIG. 64, repeater A (6203A) andrepeater B (6203B) generate two different modulated signals (on whichprecoding and change of phase are performed) in frequency band X asdescribed in Embodiment C1. The two modulated signals are respectivelytransmitted by antennas 6210A and 6212A of repeater A (6203) from FIG.62 and by antennas 6210B and 6212B of repeater B (6203B) from FIG. 62.

As for frequency band Y, repeater A (6203A) operates a processor 6500pertaining to frequency band Y and corresponding to the signal processor6500 pertaining to frequency band X shown in FIG. 65 (the signalprocessor 6500 is the signal processor pertaining to frequency band X,but given that an identical signal processor is incorporated forfrequency band Y, this description uses the same reference numbers),interleaver 304A, mapper 306A, weighting unit 308A, and phase changer5201 to generate modulated signal 5202. A transmit signal correspondingto modulated signal 5202 is then transmitted by antenna 1301A from FIG.13, that is, by antenna 6210A from FIG. 62. Similarly, repeater B (6203B) operates interleaver 304A, mapper 306A, weighting unit 308A, andphase changer 5201 from FIG. 62 pertaining to frequency band Y togenerate modulated signal 5202. Then, a transmit signal corresponding tomodulated signal 5202 is transmitted by antenna 1310A from FIG. 13,i.e., by antenna 6210B from FIG. 62.

As shown in FIG. 66 (FIG. 66 illustrates the frame configuration of themodulated signal transmitted by the base station, with time on thehorizontal axis and frequency on the vertical axis), the base stationtransmits transmission scheme information 6601, repeater-applied phasechange information 6602, and data symbols 6603. The repeater obtains andapplies the transmission scheme information 6601, the repeater-appliedphase change information 6602, and the data symbols 6603 to the transmitsignal, thus determining the phase changing scheme. When therepeater-applied phase change information 6602 from FIG. 66 is notincluded in the signal transmitted by the base station, then as shown inFIG. 62, repeater B (6203B) is the master and indicates the phasechanging scheme to repeater A (6203A).

As explained above, when the repeater transmits different data, theprecoding matrix and phase changing scheme are set according to thetransmission scheme to generate modulated signals.

On the other hand, to transmit identical data, two repeatersrespectively generate and transmit modulated signals. In suchcircumstances, repeaters each generating modulated signals fortransmission from a common antenna may be considered to be two combinedrepeaters using the precoding matrix given by formula 52. The phasechanging scheme is as explained in Embodiment C1, for example, andsatisfies the conditions of formula 53.

Also, as explained in Embodiment C1 for frequency band X, the basestation and repeater may each have two antennas that transmit respectivemodulated signals and two antennas that receive identical data. Theoperations of such a base station or repeater are as described forEmbodiment C1.

According to the present Embodiment, not only can the reception deviceobtain improved data reception quality for identical data transmissionas well as different data transmission, but the transmission devices canalso share a phase changer.

Furthermore, although the present Embodiment discusses examples usingOFDM as the transmission scheme, the invention is not limited in thismanner. Multi-carrier schemes other than OFDM and single-carrier schemesmay all be used to achieve similar Embodiments. Here, spread-spectrumcommunications may also be used. When single-carrier schemes are used,the change of phase is performed with respect to the time domain.

As explained in Embodiment 3, when the transmission scheme involvesdifferent data transmission, the change of phase is carried out on thedata symbols, only. However, as described in the present Embodiment,when the transmission scheme involves identical data transmission, thenthe change of phase need not be limited to the data symbols but may alsobe performed on pilot symbols, control symbols, and other such symbolsinserted into the transmission frame of the transmit signal. (The changeof phase need not always be performed on symbols such as pilot symbolsand control symbols, though doing so is preferable in order to achievediversity gain.)

Embodiment C4

The present Embodiment concerns a phase changing scheme different fromthe phase changing schemes described in Embodiment 1 and in theSupplement.

In Embodiment 1, formula 36 is given as an example of a precodingmatrix, and in the Supplement, formula 50 is similarly given as anothersuch example. In Embodiment A1, the phase changers from FIGS. 3, 4, 6,12, 25, 29, 51, and 53 are indicated as having a phase changing value ofPHASE[i] (where i=0, 1, 2 . . . N−2, N−1 (i denotes an integer thatsatisfies 0≤i≤N−1)) to achieve a period (cycle) of N (value reachedgiven that FIGS. 3, 4, 6, 12, 25, 29, 51, and 53 perform the change ofphase on only one baseband signal). The present description discussesperforming a change of phase on one precoded baseband signal (i.e., inFIGS. 3, 4, 6, 12, 25, 29, and 51) namely on precoded baseband signalz2′. Here, PHASE[k] is calculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 54} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {\frac{k\;\pi}{N}\mspace{14mu}{radians}}} & \left( {{formula}\mspace{14mu} 54} \right)\end{matrix}$where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≤k≤N−1).

Accordingly, the reception device is able to achieve improvements indata reception quality in the LOS environment, and especially in a radiowave propagation environment. In the LOS environment, when the change ofphase has not been performed, a regular phase relationship holds.However, when the change of phase is performed, the phase relationshipis modified, in turn avoiding poor conditions in a burst-likepropagation environment. As an alternative to formula 54, PHASE[k] maybe calculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 55} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {{- \frac{k\;\pi}{N}}\mspace{14mu}{radians}}} & \left( {{formula}\mspace{14mu} 55} \right)\end{matrix}$where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≤k≤N−1).

As a further alternative phase changing scheme, PHASE[k] may becalculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 56} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {\frac{k\;\pi}{N} + {Z\mspace{14mu}{radians}}}} & \left( {{formula}\mspace{14mu} 56} \right)\end{matrix}$where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≤k≤N−1), and Z is a fixed value.

As a further alternative phase changing scheme, PHASE[k] may becalculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 57} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {{- \frac{k\;\pi}{N}} + {Z\mspace{14mu}{radians}}}} & \left( {{formula}\mspace{14mu} 57} \right)\end{matrix}$where k=0, 1, 2 . . . N−2, N−1 (k denotes an integer that satisfies0≤k≤N−1), and Z is a fixed value.

As such, by performing the change of phase according to the presentEmbodiment, the reception device is made more likely to obtain goodreception quality.

The change of phase of the present Embodiment is applicable not only tosingle-carrier schemes but also to multi-carrier schemes. Accordingly,the present Embodiment may also be realized using, for example,spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, wavelet OFDM asdescribed in Non-Patent Literature 7, and so on. As previouslydescribed, while the present Embodiment explains the change of phase bychanging the phase with respect to the time domain t, the phase mayalternatively be changed with respect to the frequency domain asdescribed in Embodiment 1. That is, considering the change of phase inthe time domain t described in the present Embodiment and replacing twith f (f being the ((sub-) carrier) frequency) leads to a change ofphase applicable to the frequency domain. Also, as explained above forEmbodiment 1, the phase changing scheme of the present Embodiment isalso applicable to a change of phase in both the time domain and thefrequency domain. Further, when the phase changing scheme described inthe present Embodiment satisfies the conditions indicated in EmbodimentA1, the reception device is highly likely to obtain good data quality.

Embodiment C5

The present Embodiment concerns a phase changing scheme different fromthe phase changing schemes described in Embodiment 1, in the Supplement,and in Embodiment C4.

In Embodiment 1, formula 36 is given as an example of a precodingmatrix, and in the Supplement, formula 50 is similarly given as anothersuch example. In Embodiment A1, the phase changers from FIGS. 3, 4, 6,12, 25, 29, 51, and 53 are indicated as having a phase changing value ofPHASE[i] (where i=0, 1, 2 . . . N−2, N−1 (i denotes an integer thatsatisfies 0≤i≤N−1)) to achieve a period (cycle) of N (value reachedgiven that FIGS. 3, 4, 6, 12, 25, 29, 51, and 53 perform the change ofphase on only one baseband signal). The present description discussesperforming a change of phase on one precoded baseband signal (i.e., inFIGS. 3, 4, 6, 12, 25, 29, 51 and 53) namely on precoded baseband signalz2′.

The characteristic feature of the phase changing scheme pertaining tothe present Embodiment is the period (cycle) of N=2n+1. To achieve theperiod (cycle) of N=2n+1, n+1 different phase changing values areprepared. Among these n+1 different phase changing values, n phasechanging values are used twice per period (cycle), and one phasechanging value is used only once per period (cycle), thus achieving theperiod (cycle) of N=2n+1. The following describes these phase changingvalues in detail.

The n+1 different phase changing values required to achieve a phasechanging scheme in which the phase changing value is regularly switchedin a period (cycle) of N=2n+1 are expressed as PHASE[0], PHASE[1],PHASE[i] . . . PHASE[n−1], PHASE[n] (where i=0, 1, 2 . . . n−2, n−1, n(i denotes an integer that satisfies 0≤i≤n)). Here, the n+1 differentphase changing values of PHASE[0], PHASE[1], PHASE[i] . . . PHASE[n−1],PHASE[n] are expressed as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 58} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {\frac{2\; k\;\pi}{{2\; n} + 1}\mspace{14mu}{radians}}} & \left( {{formula}\mspace{14mu} 58} \right)\end{matrix}$

where k=0, 1, 2 . . . n−2, n−1, n (k denotes an integer that satisfies0≤k≤n). The n+1 different phase changing values PHASE[0], PHASE[1] . . .PHASE[i] . . . PHASE[n−1], PHASE[n] are given by formula 58. PHASE[0] isused once, while PHASE[1] through PHASE[n] are each used twice (i.e.,PHASE[1] is used twice, PHASE[2] is used twice, and so on, untilPHASE[n−1] is used twice and PHASE[n] is used twice). As such, throughthis phase changing scheme in which the phase changing value isregularly switched in a period (cycle) of N=2n+1, a phase changingscheme is realized in which the phase changing value is regularlyswitched between fewer phase changing values. Thus, the reception deviceis able to achieve better data reception quality. As the phase changingvalues are fewer, the effect thereof on the transmission device andreception device may be reduced. According to the above, the receptiondevice is able to achieve improvements in data reception quality in theLOS environment, and especially in a radio wave propagation environment.In the LOS environment, when the change of phase has not been performed,a regular phase relationship occurs. However, when the change of phaseis performed, the phase relationship is modified, in turn avoiding poorconditions in a burst-like propagation environment. As an alternative toformula 54, PHASE[k] may be calculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 59} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {{- \frac{2k\;\pi}{{2n} + 1}}\mspace{14mu}{radians}}} & \left( {{formula}\mspace{14mu} 59} \right)\end{matrix}$where k=0, 1, 2 . . . n−2, n−1, n (k denotes an integer that satisfies0≤k≤n).

The n+1 different phase changing values PHASE[0], PHASE[1] . . .PHASE[i] . . . PHASE[n−1], PHASE[n] are given by formula 59. PHASE[0] isused once, while PHASE[1] through PHASE[n] are each used twice (i.e.,PHASE[1] is used twice, PHASE[2] is used twice, and so on, untilPHASE[n−1] is used twice and PHASE[n] is used twice). As such, throughthis phase changing scheme in which the phase changing value isregularly switched in a period (cycle) of N=2n+1, a phase changingscheme is realized in which the phase changing value is regularlyswitched between fewer phase changing values. Thus, the reception deviceis able to achieve better data reception quality. As the phase changingvalues are fewer, the effect thereof on the transmission device andreception device may be reduced.

As a further alternative, PHASE[k] may be calculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 60} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {{- \frac{2k\;\pi}{{2n} + 1}} + {Z\mspace{14mu}{radians}}}} & \left( {{formula}\mspace{14mu} 60} \right)\end{matrix}$where k=0, 1, 2 . . . n−2, n−1, n (k denotes an integer that satisfies0≤k≤n) and Z is a fixed value.

The n+1 different phase changing values PHASE[0], PHASE[1] . . .PHASE[i] . . . PHASE[n−1], PHASE[n] are given by formula 60. PHASE[0] isused once, while PHASE[1] through PHASE[n] are each used twice (i.e.,PHASE[1] is used twice, PHASE[2] is used twice, and so on, untilPHASE[n−1] is used twice and PHASE[n] is used twice). As such, throughthis phase changing scheme in which the phase changing value isregularly switched in a period (cycle) of N=2n+1, a phase changingscheme is realized in which the phase changing value is regularlyswitched between fewer phase changing values. Thus, the reception deviceis able to achieve better data reception quality. As the phase changingvalues are fewer, the effect thereof on the transmission device andreception device may be reduced.

As a further alternative, PHASE[k] may be calculated as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 61} \right\rbrack & \; \\{{{PHASE}\mspace{14mu}\lbrack k\rbrack} = {{- \frac{2k\;\pi}{{2n} + 1}} + {Z\mspace{14mu}{radians}}}} & \left( {{formula}\mspace{14mu} 61} \right)\end{matrix}$where k=0, 1, 2 . . . n−2, n−1, n(k denotes an integer that satisfies0≤k≤n) and Z is a fixed value.

The n+1 different phase changing values PHASE[0], PHASE[1] . . .PHASE[i] . . . PHASE[n−1], PHASE[n] are given by formula 61. PHASE[0] isused once, while PHASE[1] through PHASE[n] are each used twice (i.e.,PHASE[1] is used twice, PHASE[2] is used twice, and so on, untilPHASE[n−1] is used twice and PHASE[n] is used twice). As such, throughthis phase changing scheme in which the phase changing value isregularly switched in a period (cycle) of N=2n+1, a phase changingscheme is realized in which the phase changing value is regularlyswitched between fewer phase changing values. Thus, the reception deviceis able to achieve better data reception quality. As the phase changingvalues are smaller, the effect thereof on the transmission device andreception device may be reduced.

As such, by performing the change of phase according to the presentEmbodiment, the reception device is made more likely to obtain goodreception quality.

The change of phase of the present Embodiment is applicable not only tosingle-carrier schemes but also to transmission using multi-carrierschemes. Accordingly, the present Embodiment may also be realized using,for example, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM,wavelet OFDM as described in Non-Patent Literature 7, and so on. Aspreviously described, while the present Embodiment explains the changeof phase as a change of phase with respect to the time domain t, thephase may alternatively be changed with respect to the frequency domainas described in Embodiment 1. That is, considering the change of phasewith respect to the time domain t described in the present Embodimentand replacing t with f (f being the ((sub-) carrier) frequency) leads toa change of phase applicable to the frequency domain. Also, as explainedabove for Embodiment 1, the phase changing scheme of the presentEmbodiment is also applicable to a change of phase with respect to boththe time domain and the frequency domain.

Embodiment C6

The present Embodiment describes a scheme for regularly changing thephase, specifically that of Embodiment C5, when encoding is performedusing block codes as described in Non-Patent Literature 12 through 15,such as QC LDPC Codes (not only QC-LDPC but also LDPC codes may beused), concatenated LDPC (blocks) and BCH codes, Turbo codes orDuo-Binary Turbo Codes using tail-biting, and so on. The followingexample considers a case where two streams s1 and s2 are transmitted.When encoding has been performed using block codes and controlinformation and the like is not necessary, the number of bits making upeach coded block matches the number of bits making up each block code(control information and so on described below may yet be included).When encoding has been performed using block codes or the like andcontrol information or the like (e.g., CRC transmission parameters) isrequired, then the number of bits making up each coded block is the sumof the number of bits making up the block codes and the number of bitsmaking up the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed intwo coded blocks when block codes are used. FIG. 34 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 4, and thetransmission device has only one encoder. (Here, the transmission schememay be any single-carrier scheme or multi-carrier scheme such as OFDM.)

As shown in FIG. 34, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

Then, given that the transmission device from FIG. 4 transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up one coded block, andwhen the modulation scheme is 64-QAM, 500 slots are needed to transmitall of the bits making up one coded block.

The following describes the relationship between the above-defined slotsand the phase, as pertains to schemes for a regular change of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase, which has a period (cycle) of five. That is, the phasechanger of the transmission device from FIG. 4 uses five phase changingvalues (or phase changing sets) to achieve the period (cycle) of five.However, as described in Embodiment C5, three different phase changingvalues are present. Accordingly, some of the five phase changing valuesneeded for the period (cycle) of five are identical. (As in FIG. 6, fivephase changing values are needed in order to perform a change of phasehaving a period (cycle) of five on precoded baseband signal z2′ only.Also, as in FIG. 26, two phase changing values are needed for each slotin order to perform the change of phase on both precoded basebandsignals z1′ and z2′. These two phase changing values are termed a phasechanging set. Accordingly, five phase changing sets should ideally beprepared in order to perform a change of phase having a period (cycle)of five in such circumstances). The five phase changing values (or phasechanging sets) needed for the period (cycle) of five are expressed asP[0], P[1], P[2], P[3], and P[4].

The following describes the relationship between the above-defined slotsand the phase, as pertains to schemes for a regular change of phase.

For the above-described 1500 slots needed to transmit the 6000 bitsmaking up a single coded block when the modulation scheme is QPSK, phasechanging value P[0] is used on 300 slots, phase changing value P[1] isused on 300 slots, phase changing value P[2] is used on 300 slots, phasechanging value P[3] is used on 300 slots, and phase changing value P[4]is used on 300 slots. This is due to the fact that any bias in phasechanging value usage causes great influence to be exerted by the morefrequently used phase changing value, and that the reception device isdependent on such influence for data reception quality.

Similarly, for the above-described 750 slots needed to transmit the 6000bits making up a single coded block when the modulation scheme is16-QAM, phase changing value P[0] is used on 150 slots, phase changingvalue P[1] is used on 150 slots, phase changing value P[2] is used on150 slots, phase changing value P[3] is used on 150 slots, and phasechanging value P[4] is used on 150 slots.

Furthermore, for the above-described 500 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is64-QAM, phase changing value P[0] is used on 100 slots, phase changingvalue P[1] is used on 100 slots, phase changing value P[2] is used on100 slots, phase changing value P[3] is used on 100 slots, and phasechanging value P[4] is used on 100 slots.

As described above, a phase changing scheme for a regular change ofphase changing value as given in Embodiment C5 requires the preparationof N=2n+1 phase changing values P[0], P[1] . . . P[2n−1], P[2n] (whereP[0], P[1] . . . P[2n−1], P[2n] are expressed as PHASE[0], PHASE[1],PHASE[2] . . . PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, inorder to transmit all of the bits making up a single coded block, phasechanging value P[0] is used on K₀ slots, phase changing value P[1] isused on K₁ slots, phase changing value P[i] is used on K_(i) slots(where i=0, 1, 2 . . . 2n−1, 2n (i denotes an integer that satisfies0≤i≤2n)), and phase changing value P[2n] is used on K_(2n) slots, suchthat Condition #C01 is met.

(Condition #C01)

K₀=K₁ . . . =K_(i)= . . . K_(2n). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . 2n−1, 2n (a denotes an integer that satisfies0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).

A phase changing scheme for a regular change of phase changing value asgiven in Embodiment C5 having a period (cycle) of N=2n+1 requires thepreparation of phase changing values PHASE[0], PHASE[1], PHASE[2] . . .PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bitsmaking up a single coded block, phase changing value PHASE[0] is used onG₀ slots, phase changing value PHASE[1] is used on G₁ slots, phasechanging value PHASE[i] is used on G_(i) slots (where i=0, 1, 2 . . .n−1, n (i denotes an integer that satisfies 0≤i≤n), and phase changingvalue PHASE[n] is used on G_(n) slots, such that Condition #C01 is met.Condition #C01 may be modified as follows.

(Condition #C02)

2×G₀=G₁ . . . =G_(i)= . . . G_(n). That is, 2×G₀=G_(a) (∀a where a=1, 2. . . n−1, n (a denotes an integer that satisfies 1≤a≤n)).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #C01 (orCondition #C02) should preferably be met for the supported modulationscheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #C01 (or Condition #C02) may not be satisfied for somemodulation schemes. In such a case, the following condition appliesinstead of Condition #C01.

(Condition #C03)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b) satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1, 2n(a denotes an integer that satisfies 0≤a≤2n, b denotes an integer thatsatisfies 0≤b≤2n) a≠b).

Alternatively, Condition #C03 may be expressed as follows.

(Condition #C04)

The difference between G_(a) and G_(b) satisfies 0, 1, or 2. That is,|G_(a)−G_(b)| satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . . . n−1, n(a denotes an integer that satisfies 1≤a≤n, b denotes an integer thatsatisfies 1≤b≤n), a≠b)

and

The difference between 2×G₀ and G_(a) satisfies 0, 1, or 2. That is,|2×G₀−G_(a)| satisfies 0, 1, or 2 (∀a, where a=1, 2 . . . n−1, n (adenotes an integer that satisfies 1≤a≤n)).

FIG. 35 illustrates the varying numbers of symbols and slots needed intwo coded blocks when block codes are used. FIG. 35 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 3 and FIG.12, and the transmission device has two encoders. (Here, thetransmission scheme may be any single-carrier scheme or multi-carrierscheme such as OFDM.)

As shown in FIG. 35, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM.

The transmission device from FIG. 3 and the transmission device fromFIG. 12 each transmit two streams at once, and have two encoders. Assuch, the two streams each transmit different code blocks. Accordingly,when the modulation scheme is QPSK, two coded blocks drawn from s1 ands2 are transmitted within the same interval, e.g., a first coded blockdrawn from s1 is transmitted, then a second coded block drawn from s2 istransmitted. As such, 3000 slots are needed in order to transmit thefirst and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 slotsare needed to transmit all of the bits making up one coded block, andwhen the modulation scheme is 64-QAM, 1000 slots are needed to transmitall of the bits making up one coded block.

The following describes the relationship between the above-defined slotsand the phase, as pertains to schemes for a regular change of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase, which has a period (cycle) of five. That is, the phasechanger of the transmission device from FIG. 4 uses five phase changingvalues (or phase changing sets) to achieve the period (cycle) of five.However, as described in Embodiment C5, three different phase changingvalues are present. Accordingly, some of the five phase changing valuesneeded for the period (cycle) of five are identical. (As in FIG. 6, fivephase changing values are needed in order to perform the change of phasehaving a period (cycle) of five on precoded baseband signal z2′ only.Also, as in FIG. 26, two phase changing values are needed for each slotin order to perform the change of phase on both precoded basebandsignals z1′ and z2′. These two phase changing values are termed a phasechanging set. Accordingly, five phase changing sets should ideally beprepared in order to perform a change of phase having a period (cycle)of five in such circumstances). The five phase changing values (or phasechanging sets) needed for the period (cycle) of five are expressed asP[0], P[1], P[2], P[3], and P[4].

For the above-described 3000 slots needed to transmit the 6000×2 bitsmaking up the pair of coded blocks when the modulation scheme is QPSK,phase changing value P[0] is used on 600 slots, phase changing valueP[1] is used on 600 slots, phase changing value P[2] is used on 600slots, phase changing value P[3] is used on 6100 slots, and phasechanging value P[4] is used on 600 slots. This is due to the fact thatany bias in phase changing value usage causes great influence to beexerted by the more frequently used phase changing value, and that thereception device is dependent on such influence for data receptionquality.

Further, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 600 times, phase changing value P[1] is usedon slots 600 times, phase changing value P[2] is used on slots 600times, phase changing value P[3] is used on slots 600 times, and phasechanging value PHASE[4] is used on slots 600 times. Furthermore, inorder to transmit the second coded block, phase changing value P[0] isused on slots 600 times, phase changing value P[1] is used on slots 600times, phase changing value P[2] is used on slots 600 times, phasechanging value P[3] is used on slots 600 times, and phase changing valueP[4] is used on slots 600 times.

Similarly, for the above-described 1500 slots needed to transmit the6000×2 bits making up the pair of coded blocks when the modulationscheme is 16-QAM, phase changing value P[0] is used on 300 slots, phasechanging value P[1] is used on 300 slots, phase changing value P[2] isused on 300 slots, phase changing value P[3] is used on 300 slots, andphase changing value P[4] is used on 300 slots.

Furthermore, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 300 times, phase changing value P[1] is usedon slots 300 times, phase changing value P[2] is used on slots 300times, phase changing value P[3] is used on slots 300 times, and phasechanging value P[4] is used on slots 300 times. Furthermore, in order totransmit the second coded block, phase changing value P[0] is used onslots 300 times, phase changing value P[1] is used on slots 300 times,phase changing value P[2] is used on slots 300 times, phase changingvalue P[3] is used on slots 300 times, and phase changing value P[4] isused on slots 300 times.

Furthermore, for the above-described 1000 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is64-QAM, phase changing value P[0] is used on 200 slots, phase changingvalue P[1] is used on 200 slots, phase changing value P[2] is used on200 slots, phase changing value P[3] is used on 200 slots, and phasechanging value P[4] is used on 200 slots.

Further, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 200 times, phase changing value P[1] is usedon slots 200 times, phase changing value P[2] is used on slots 200times, phase changing value P[3] is used on slots 200 times, and phasechanging value P[4] is used on slots 200 times. Furthermore, in order totransmit the second coded block, phase changing value P[0] is used onslots 200 times, phase changing value P[1] is used on slots 200 times,phase changing value P[2] is used on slots 200 times, phase changingvalue P[3] is used on slots 200 times, and phase changing value P[4] isused on slots 200 times.

As described above, a phase changing scheme for regularly varying thephase changing value as given in Embodiment C5 requires the preparationof N=2n+1 phase changing values P[0], P[1] . . . P[2n−1], P[2n] (whereP[0], P[1] . . . P[2n−1], P[2n] are expressed as PHASE[0], PHASE[1],PHASE[2] . . . PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, inorder to transmit all of the bits making up the two coded blocks, phasechanging value P[0] is used on K₀ slots, phase changing value P[1] isused on K₁ slots, phase changing value P[i] is used on K_(i) slots(where i=0, 1, 2 . . . 2n−1, 2n (i denotes an integer that satisfies0≤i≤2n)), and phase changing value P[2n] is used on K_(2n) slots, suchthat Condition #C01 is met.

(Condition #C05)

K₀=K₁ . . . =K_(i)= . . . K_(2n). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . 2n−1, 2n (a denotes an integer that satisfies0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b). In order totransmit all of the bits making up the first coded block, phase changingvalue P[0] is used K_(0,1) times, phase changing value P[1] is usedK_(1,1) times, phase changing value P[i] is used K_(i,1) (where i=0, 1,2 . . . 2n−1, 2n (i denotes an integer that satisfies 0≤i≤2n)), andphase changing value P[2n] is used K_(2n,1) times.

(Condition #C06)

K_(0,1)=K_(1,1) . . . =K_(i,1)= . . . K_(2n,1). That is, K_(a,1)=K_(b,1)(∀a and ∀b where a, b, =0, 1, 2 . . . 2n−1, 2n (a denotes an integerthat satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n),a≠b).

In order to transmit all of the bits making up the second coded block,phase changing value P[0] is used K_(0,2) times, phase changing valueP[1] is used K_(1,2) times, phase changing value P[i] is used K_(i,2)(where i=0, 1, 2 . . . 2n−1, 2n (i denotes an integer that satisfies0≤i≤2n)), and phase changing value P[2n] is used K_(2n,2) times.

(Condition #C07)

K_(0,2)=K_(1,2) . . . =K_(i,2)= . . . K_(2n,2). That is, K_(a,2)=K_(b,2)(∀a and ∀b where a, b, =0, 1, 2 . . . 2n−1, 2n (a denotes an integerthat satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n),a≠b).

A phase changing scheme for regularly varying the phase changing valueas given in Embodiment C5 having a period (cycle) of N=2n+1 requires thepreparation of phase changing values PHASE[0], PHASE[1], PHASE[2] . . .PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bitsmaking up the two coded blocks, phase changing value PHASE[0] is used onG₀ slots, phase changing value PHASE[1] is used on G₁ slots, phasechanging value PHASE[i] is used on G_(i) slots (where i=0, 1, 2 . . .n−1, n (i denotes an integer that satisfies 0≤i≤n)), and phase changingvalue PHASE[n] is used on G_(n) slots, such that Condition #C05 is met.

(Condition #C08)

2×G₀=G₁ . . . =G_(i)= . . . G_(n). That is, 2×G₀=G_(a) (∀a where a=1, 2. . . n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes aninteger that satisfies 1≤b≤n)).

In order to transmit all of the bits making up the first coded block,phase changing value PHASE[0] is used G_(0,1) times, phase changingvalue PHASE[1] is used G₁,1 times, phase changing value PHASE[i] is usedG_(i,1) (where i=0, 1, 2 . . . n−1, n (i denotes an integer thatsatisfies 0≤i≤n)), and phase changing value PHASE[n] is used G_(n,1)times.

(Condition #C09)

2×G_(0,1)=G_(1,1) . . . =G_(i,1)= . . . G_(n,1). That is,2×G_(0,1)=G_(a,1) (∀a where a=1, 2 . . . n−1, n (a denotes an integerthat satisfies 1≤a≤n)).

In order to transmit all of the bits making up the second coded block,phase changing value PHASE[0] is used G_(0,2) times, phase changingvalue PHASE[1] is used G_(1,2) times, phase changing value PHASE[i] isused G_(i,2) (where i=0, 1, 2 . . . n−1, n (i denotes an integer thatsatisfies 0≤i≤n)), and phase changing value PHASE[n] is used G_(n,1)times.

(Condition #C10)

2×G_(0,2)=G_(1,2) . . . =G_(i,2)= . . . G_(n,2). That is,2×G_(0,2)=G_(a,2) (∀a where a=1, 2 . . . n−1, n (a denotes an integerthat satisfies 1≤a≤n)).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #C05,Condition #C06, and Condition #C07 (or Condition #C08, Condition #C09,and Condition #C10) should preferably be met for the supportedmodulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #C05, Condition #C06, and Condition #C07 (or Condition #C08,Condition #C09, and Condition #C10) may not be satisfied for somemodulation schemes. In such a case, the following conditions applyinstead of Condition #C05, Condition #C06, and Condition #C07.

(Condition #C11)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1,2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integerthat satisfies 0≤b≤2n), a≠b).

(Condition #C12)

The difference between K_(a,1) and K_(b,1) satisfies 0 or 1. That is,|K_(a,1)−K_(b,1)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . .2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes aninteger that satisfies 0≤b≤2n), a≠b).

(Condition #C13)

The difference between K_(a,2) and K_(b,2) satisfies 0 or 1. That is,|K_(a,2)−K_(b,2)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . .2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes aninteger that satisfies 0≤b≤2n), a≠b).

Alternatively, Condition #C11, Condition #C12, and Condition #C13 may beexpressed as follows.

(Condition #C14)

The difference between G_(a) and G_(b) satisfies 0, 1, or 2. That is,|G_(a)−G_(b)| satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . . . n−1, n(a denotes an integer that satisfies 1≤a≤n, b denotes an integer thatsatisfies 1≤b≤n), a≠b)

and

The difference between 2×G₀ and G_(a) satisfies 0, 1, or 2. That is,|2×G₀−G_(a)| satisfies 0, 1, or 2 (∀a, where a=1, 2 . . . n−1, n (adenotes an integer that satisfies 1≤a≤n)).

(Condition #C15)

The difference between G_(a,1) and G_(b,1) satisfies 0, 1, or 2. Thatis, |G_(a,1)−G_(b,1)| satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . .. n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes aninteger that satisfies 1≤b≤n), a≠b)

and

The difference between 2×G_(0,1) and G_(a,1) satisfies 0, 1, or 2. Thatis, |2×G_(0,1)−G_(a,1)| satisfies 0, 1, or 2 (∀a, where a=1, 2 . . .n−1, n (a denotes an integer that satisfies 1≤a≤n)).

(Condition #C16)

The difference between G_(a,2) and G_(b,2) satisfies 0, 1, or 2. Thatis, |G_(a,2)−G_(b,2)| satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . .. n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes aninteger that satisfies 1≤b≤n), a≠b)

and

The difference between 2×G_(0,2) and G_(a,2) satisfies 0, 1, or 2. Thatis, |2×G_(0,2)−G_(a,2)| satisfies 0, 1, or 2 (∀a, where a=1, 2 . . .n−1, n (a denotes an integer that satisfies 1≤a≤n)).

As described above, bias among the phase changing values being used totransmit the coded blocks is removed by creating a relationship betweenthe coded block and the phase changing values. As such, data receptionquality can be improved for the reception device.

In the present Embodiment, N phase changing values (or phase changingsets) are needed in order to perform the change of phase having a period(cycle) of N with a regular phase changing scheme. As such, N phasechanging values (or phase changing sets) P[0], P[1], P[2] . . . P[N−2],and P[N−1] are prepared. However, schemes exist for ordering the phasesin the stated order with respect to the frequency domain. No limitationis intended in this regard. The N phase changing values (or phasechanging sets) P[0], P[1], P[2] . . . P[N−2], and P[N−1] may also changethe phases of blocks in the time domain or in the time-frequency domainto obtain a symbol arrangement as described in Embodiment 1. Althoughthe above examples discuss a phase changing scheme with a period (cycle)of N, the same effects are obtainable using N phase changing values (orphase changing sets) at random. That is, the N phase changing values (orphase changing sets) need not always have regular periodicity. As longas the above-described conditions are satisfied, quality data receptionimprovements are realizable for the reception device.

Furthermore, given the existence of modes for spatial multiplexing MIMOschemes, MIMO schemes using a fixed precoding matrix, space-time blockcoding schemes, single-stream transmission, and schemes using a regularchange of phase, the transmission device (broadcaster, base station) mayselect any one of these transmission schemes.

As described in Non-Patent Literature 3, spatial multiplexing MIMOschemes involve transmitting signals s1 and s2, which are mapped using aselected modulation scheme, on each of two different antennas. MIMOschemes using a fixed precoding matrix involve performing precoding only(with no change of phase). Further, space-time block coding schemes aredescribed in Non-Patent Literature 9, 16, and 17. Single-streamtransmission schemes involve transmitting signal s1, mapped with aselected modulation scheme, from an antenna after performingpredetermined processing.

Schemes using multi-carrier transmission such as OFDM involve a firstcarrier group made up of a plurality of carriers and a second carriergroup made up of a plurality of carriers different from the firstcarrier group, and so on, such that multi-carrier transmission isrealized with a plurality of carrier groups. For each carrier group, anyof spatial multiplexing MIMO schemes, MIMO schemes using a fixedprecoding matrix, space-time block coding schemes, single-streamtransmission, and schemes using a regular change of phase may be used.In particular, schemes using a regular change of phase on a selected(sub-)carrier group are preferably used to realize the presentEmbodiment.

When a change of phase by, for example, a phase changing value for P[i]of X radians is performed on only one precoded baseband signal, thephase changers from FIGS. 3, 4, 6, 12, 25, 29, 51, and 53 multiplyprecoded baseband signal z2′ by e^(jX). Then, when a change of phase by,for example, a phase changing set for P[i] of X radians and Y radians isperformed on both precoded baseband signals, the phase changers fromFIGS. 26, 27, 28, 52, and 54 multiply precoded baseband signal z2′ bye^(jX) and multiply precoded baseband signal z1′ by e^(jY).

Embodiment C7

The present Embodiment describes a scheme for regularly changing thephase, specifically as done in Embodiment A1 and Embodiment C6, whenencoding is performed using block codes as described in Non-PatentLiterature 12 through 15, such as QC LDPC Codes (not only QC-LDPC butalso LDPC (block) codes may be used), concatenated LDPC and BCH codes,Turbo codes or Duo-Binary Turbo Codes, and so on. The following exampleconsiders a case where two streams s1 and s2 are transmitted. Whenencoding has been performed using block codes and control informationand the like is not necessary, the number of bits making up each codedblock matches the number of bits making up each block code (controlinformation and so on described below may yet be included). Whenencoding has been performed using block codes or the like and controlinformation or the like (e.g., CRC transmission parameters) is required,then the number of bits making up each coded block is the sum of thenumber of bits making up the block codes and the number of bits makingup the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed inone coded block when block codes are used. FIG. 34 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 4, and thetransmission device has only one encoder. (Here, the transmission schememay be any single-carrier scheme or multi-carrier scheme such as OFDM.)As shown in FIG. 34, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

Then, given that the transmission device from FIG. 4 transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols (hereinafter, slots) are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up one coded block, andwhen the modulation scheme is 64-QAM, 500 slots are needed to transmitall of the bits making up one coded block.

The following describes the relationship between the above-defined slotsand the phase, as pertains to schemes for a regular change of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase, which has a period (cycle) of five. The phase changingvalues (or phase changing sets) prepared in order to regularly changethe phase with a period (cycle) of five are P[0], P[1], P[2], P[3], andP[4]. However, P[0], P[1], P[2], P[3], and P[4] should include at leasttwo different phase changing values (i.e., P[0], P[1], P[2], P[3], andP[4] may include identical phase changing values). (As in FIG. 6, fivephase changing values are needed in order to perform a change of phasehaving a period (cycle) of five on precoded baseband signal z2′ only.Also, as in FIG. 26, two phase changing values are needed for each slotin order to perform the change of phase on both precoded basebandsignals z1′ and z2′. These two phase changing values are termed a phasechanging set. Accordingly, five phase changing sets should ideally beprepared in order to perform a change of phase having a period (cycle)of five in such circumstances).

For the above-described 1500 slots needed to transmit the 6000 bitsmaking up a single coded block when the modulation scheme is QPSK, phasechanging value P[0] is used on 300 slots, phase changing value P[1] isused on 300 slots, phase changing value P[2] is used on 300 slots, phasechanging value P[3] is used on 300 slots, and phase changing value P[4]is used on 300 slots. This is due to the fact that any bias in phasechanging value usage causes great influence to be exerted by the morefrequently used phase changing value, and that the reception device isdependent on such influence for data reception quality.

Furthermore, for the above-described 750 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is16-QAM, phase changing value P[0] is used on 150 slots, phase changingvalue P[1] is used on 150 slots, phase changing value P[2] is used on150 slots, phase changing value P[3] is used on 150 slots, and phasechanging value P[4] is used on 150 slots.

Further, for the above-described 500 slots needed to transmit the 6000bits making up a single coded block when the modulation scheme is64-QAM, phase changing value P[0] is used on 100 slots, phase changingvalue P[1] is used on 100 slots, phase changing value P[2] is used on100 slots, phase changing value P[3] is used on 100 slots, and phasechanging value P[4] is used on 100 slots.

As described above, the phase changing values used in the phase changingscheme regularly switching between phase changing values with a period(cycle) of N are expressed as P[0], P[1] . . . P[N−2], P[N−1]. However,P[0], P[1] . . . P[N−2], P[N−1] should include at least two differentphase changing values (i.e., P[0], P[1] . . . P[N−2], P[N−1] may includeidentical phase changing values). In order to transmit all of the bitsmaking up a single coded block, phase changing value P[0] is used on K₀slots, phase changing value P[1] is used on K₁ slots, phase changingvalue P[i] is used on K_(i) slots (where i=0, 1, 2 . . . N−1 (i denotesan integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] isused on K_(N−1) slots, such that Condition #C17 is met.

(Condition #C17)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≤a≤N−1, bdenotes an integer that satisfies 0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #C17 shouldpreferably be met for the supported modulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #C17 may not be satisfied for some modulation schemes. In sucha case, the following condition applies instead of Condition #C17.

(Condition #C18)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b).

FIG. 35 illustrates the varying numbers of symbols and slots needed intwo coded block when block codes are used. FIG. 35 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 3 and FIG.12, and the transmission device has two encoders. (Here, thetransmission scheme may be any single-carrier scheme or multi-carrierscheme such as OFDM.)

As shown in FIG. 35, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for64-QAM.

The transmission device from FIG. 3 and the transmission device fromFIG. 12 each transmit two streams at once, and have two encoders. Assuch, the two streams each transmit different code blocks. Accordingly,when the modulation scheme is QPSK, two coded blocks drawn from s1 ands2 are transmitted within the same interval, e.g., a first coded blockdrawn from s1 is transmitted, then a second coded block drawn from s2 istransmitted. As such, 3000 slots are needed in order to transmit thefirst and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 slotsare needed to transmit all of the bits making up one coded block, andwhen the modulation scheme is 64-QAM, 1000 slots are needed to transmitall of the bits making up one coded block.

The following describes the relationship between the above-defined slotsand the phase, as pertains to schemes for a regular change of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase, which has a period (cycle) of five. That is, the phasechanger of the transmission device from FIG. 4 uses five phase changingvalues (or phase changing sets) P[0], P[1], P[2], P[3], and P[4] toachieve the period (cycle) of five. However, P[0], P[1], P[2], P[3], andP[4] should include at least two different phase changing values (i.e.,P[0], P[1], P[2], P[3], and P[4] may include identical phase changingvalues). (As in FIG. 6, five phase changing values are needed in orderto perform a change of phase having a period (cycle) of five on precodedbaseband signal z2′ only. Also, as in FIG. 26, two phase changing valuesare needed for each slot in order to perform the change of phase on bothprecoded baseband signals z1′ and z2′. These two phase changing valuesare termed a phase changing set. Accordingly, five phase changing setsshould ideally be prepared in order to perform a change of phase havinga period (cycle) of five in such circumstances). The five phase changingvalues (or phase changing sets) needed for the period (cycle) of fiveare expressed as P[0], P[1], P[2], P[3], and P[4].

For the above-described 3000 slots needed to transmit the 6000×2 bitsmaking up the pair of coded blocks when the modulation scheme is QPSK,phase changing value P[0] is used on 600 slots, phase changing valueP[1] is used on 600 slots, phase changing value P[2] is used on 600slots, phase changing value P[3] is used on 600 slots, and phasechanging value P[4] is used on 600 slots. This is due to the fact thatany bias in phase changing value usage causes great influence to beexerted by the more frequently used phase changing value, and that thereception device is dependent on such influence for data receptionquality.

Further, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 600 times, phase changing value P[1] is usedon slots 600 times, phase changing value P[2] is used on slots 600times, phase changing value P[3] is used on slots 600 times, and phasechanging value P[4] is used on slots 600 times. Furthermore, in order totransmit the second coded block, phase changing value P[0] is used onslots 600 times, phase changing value P[1] is used on slots 600 times,phase changing value P[2] is used on slots 600 times, phase changingvalue P[3] is used on slots 600 times, and phase changing value P[4] isused on slots 600 times.

Similarly, for the above-described 1500 slots needed to transmit the6000×2 bits making up the pair of coded blocks when the modulationscheme is 16-QAM, phase changing value P[0] is used on 300 slots, phasechanging value P[1] is used on 300 slots, phase changing value P[2] isused on 300 slots, phase changing value P[3] is used on 300 slots, andphase changing value P[4] is used on 300 slots.

Further, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 300 times, phase changing value P[1] is usedon slots 300 times, phase changing value P[2] is used on slots 300times, phase changing value P[3] is used on slots 300 times, and phasechanging value P[4] is used on slots 300 times. Furthermore, in order totransmit the second coded block, phase changing value P[0] is used onslots 300 times, phase changing value P[1] is used on slots 300 times,phase changing value P[2] is used on slots 300 times, phase changingvalue P[3] is used on slots 300 times, and phase changing value P[4] isused on slots 300 times.

Similarly, for the above-described 1000 slots needed to transmit the6000×2 bits making up the pair of coded blocks when the modulationscheme is 64-QAM, phase changing value P[0] is used on 200 slots, phasechanging value P[1] is used on 200 slots, phase changing value P[2] isused on 200 slots, phase changing value P[3] is used on 200 slots, andphase changing value P[4] is used on 200 slots.

Further, in order to transmit the first coded block, phase changingvalue P[0] is used on slots 200 times, phase changing value P[1] is usedon slots 200 times, phase changing value P[2] is used on slots 200times, phase changing value P[3] is used on slots 200 times, and phasechanging value P[4] is used on slots 200 times. Furthermore, in order totransmit the second coded block, phase changing value P[0] is used onslots 200 times, phase changing value P[1] is used on slots 200 times,phase changing value P[2] is used on slots 200 times, phase changingvalue P[3] is used on slots 200 times, and phase changing value P[4] isused on slots 200 times.

As described above, the phase changing values used in the phase changingscheme regularly switching between phase changing values with a period(cycle) of N are expressed as P[0], P[1] . . . P[N−2], P[N−1]. However,P[0], P[1] . . . P[N−2], P[N−1] should include at least two differentphase changing values (i.e., P[0], P[1] . . . P[N−2], P[N−1] may includeidentical phase changing values). In order to transmit all of the bitsmaking up two coded blocks, phase changing value P[0] is used on K₀slots, phase changing value P[1] is used on K₁ slots, phase changingvalue P[i] is used on K_(i) slots (where i=0, 1, 2 . . . N−1 (i denotesan integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] isused on K_(N−1) slots, such that Condition #C19 is met.

(Condition #C19)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (∀a and ∀b wherea, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies 0≤a≤N−1, bdenotes an integer that satisfies 0≤b≤N−1), a≠b).

In order to transmit all of the bits making up the first coded block,phase changing value P[0] is used K_(0,1) times, phase changing valueP[1] is used K₁,1 times, phase changing value P[i] is used K_(i,1)(where i=0, 1, 2 . . . N−1 (i denotes an integer that satisfies0≤i≤N−1)), and phase changing value P[N−1] is used K_(N−1,1) times.

(Condition #C20)

K_(0,1)=K_(1,1)= . . . K_(i,1)= . . . K_(N−1,1). That is,K_(a,1)=K_(b,1) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

In order to transmit all of the bits making up the second coded block,phase changing value P[0] is used K_(0,2) times, phase changing valueP[1] is used K_(1,2) times, phase changing value P[i] is used K_(i,2)(where i=0, 1, 2 . . . N−1(i denotes an integer that satisfies0≤i≤N−1)), and phase changing value P[N−1] is used K_(N−1,2) times.

(Condition #C21)

K_(0,2)=K_(1,2)= . . . K_(i,2)= . . . K_(N−1,2). That is,K_(a,2)=K_(b,2) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #C19,Condition #C20, and Condition #C21 are preferably met for the supportedmodulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #C19, Condition #C20, and Condition #C21 may not be satisfiedfor some modulation schemes. In such a case, the following conditionsapply instead of Condition #C19, Condition #C20, and Condition #C21.

(Condition #C22)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b).

(Condition #C23)

The difference between K_(a,1) and K_(b,1) satisfies 0 or 1. That is,|K_(a,1)−K_(b,1)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b).

(Condition #C24)

The difference between K_(a,2) and K_(b,2) satisfies 0 or 1. That is,|K_(a,2)−K_(b,2)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b).

As described above, bias among the phase changing values being used totransmit the coded blocks is removed by creating a relationship betweenthe coded block and the phase changing values. As such, data receptionquality can be improved for the reception device.

In the present Embodiment, N phase changing values (or phase changingsets) are needed in order to perform a change of phase having a period(cycle) of N with the scheme for a regular change of phase. As such, Nphase changing values (or phase changing sets) P[0], P[1], P[2] . . .P[N−2], and P[N−1] are prepared. However, schemes exist for ordering thephases in the stated order with respect to the frequency domain. Nolimitation is intended in this regard. The N phase changing values (orphase changing sets) P[0], P[1], P[2] . . . P[N−2], and P[N−1] may alsochange the phases of blocks in the time domain or in the time-frequencydomain to obtain a symbol arrangement as described in Embodiment 1.Although the above examples discuss a phase changing scheme with aperiod (cycle) of N, the same effects are obtainable using N phasechanging values (or phase changing sets) at random. That is, the N phasechanging values (or phase changing sets) need not always have regularperiodicity. As long as the above-described conditions are satisfied,great quality data reception improvements are realizable for thereception device.

Furthermore, given the existence of modes for spatial multiplexing MIMOschemes, MIMO schemes using a fixed precoding matrix, space-time blockcoding schemes, single-stream transmission, and schemes using a regularchange of phase, the transmission device (broadcaster, base station) mayselect any one of these transmission schemes.

As described in Non-Patent Literature 3, spatial multiplexing MIMOschemes involve transmitting signals s1 and s2, which are mapped using aselected modulation scheme, on each of two different antennas. MIMOschemes using a fixed precoding matrix involve performing precoding only(with no change of phase). Further, space-time block coding schemes aredescribed in Non-Patent Literature 9, 16, and 17. Single-streamtransmission schemes involve transmitting signal s1, mapped with aselected modulation scheme, from an antenna after performingpredetermined processing.

Schemes using multi-carrier transmission such as OFDM involve a firstcarrier group made up of a plurality of carriers and a second carriergroup made up of a plurality of carriers different from the firstcarrier group, and so on, such that multi-carrier transmission isrealized with a plurality of carrier groups. For each carrier group, anyof spatial multiplexing MIMO schemes, MIMO schemes using a fixedprecoding matrix, space-time block coding schemes, single-streamtransmission, and schemes using a regular change of phase may be used.In particular, schemes using a regular change of phase on a selected(sub-)carrier group are preferably used to realize the presentEmbodiment.

When a change of phase by, for example, a phase changing value for P[i]of X radians is performed on only one precoded baseband signal, thephase changers of FIGS. 3, 4, 6, 12, 25, 29, 51, and 53 multiplyprecoded baseband signal z2′ by e^(jX). Then, when a change of phase by,for example, a phase changing set for P[i] of X radians and Y radians isperformed on both precoded baseband signals, the phase changers fromFIGS. 26, 27, 28, 52, and 54 multiply precoded baseband signal z2′ bye^(jX) and multiply precoded baseband signal z1′ by e^(jY).

Embodiment D1

The present Embodiment is first described as a variation ofEmbodiment 1. FIG. 67 illustrates a sample transmission devicepertaining to the present Embodiment. Components thereof operatingidentically to those of FIG. 3 use the same reference numbers thereas,and the description thereof is omitted for simplicity, below. FIG. 67differs from FIG. 3 in the insertion of a baseband signal switcher 6702directly following the weighting units. Accordingly, the followingexplanations are primarily centered on the baseband signal switcher6702.

FIG. 21 illustrates the configuration of the weighting units 308A and308B. The area of FIG. 21 enclosed in the dashed line represents one ofthe weighting units. Baseband signal 307A is multiplied by w11 to obtainw11·s1(t), and multiplied by w21 to obtain w21·s1(t). Similarly,baseband signal 307B is multiplied by w12 to obtain w12·s2(t), andmultiplied by w22 to obtain w22·s2(t). Next, z1(t)=w11·s1(t)+w12·s2(t)and z2(t)=w21·s1(t)+w22·s22(t) are obtained. Here, as explained inEmbodiment 1, s1(t) and s2(t) are baseband signals modulated accordingto a modulation scheme such as BPSK, QPSK, 8-PSK, 16-QAM, 32-QAM,64-QAM, 256-QAM, 16-APSK and so on. Both weighting units performweighting using a fixed precoding matrix. The precoding matrix uses, forexample, the scheme of formula 62, and satisfies the conditions offormula 63 or formula 64, all found below. However, this is only anexample. The value of a is not limited to formula 63 and formula 64, andmay, for example, be 1, or may be 0 (a is preferably a real numbergreater than or equal to 0, but may be also be an imaginary number).

Here, the precoding matrix is

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 62} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\; 0}} \\{\alpha \times e^{j\; 0}} & e^{j\;\pi}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 62} \right)\end{matrix}$

In formula 62, above, α is given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 63} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 4}{\sqrt{2\;} + 2}} & \left( {{formula}\mspace{14mu} 63} \right)\end{matrix}$

Alternatively, in formula 62, above, α may be given by:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 64} \right\rbrack & \; \\{\alpha = \frac{\sqrt{2} + 3 + \sqrt{5}}{\sqrt{2\;} + 3 - \sqrt{5}}} & \left( {{formula}\mspace{14mu} 64} \right)\end{matrix}$

Alternatively, the precoding matrix is not restricted to that of formula62, but may also be:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 65} \right\rbrack & \; \\{\begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix} = \begin{pmatrix}a & b \\c & d\end{pmatrix}} & \left( {{formula}\mspace{14mu} 65} \right)\end{matrix}$

where a=A^(jδ11), b=Be^(jδ12), c=Ce^(jδ21), and d=De^(jδ22). Further,one of a, b, c, and d may be equal to zero. For example: (1) a may bezero while b, c, and d are non-zero, (2) b may be zero while a, c, and dare non-zero, (3) c may be zero while a, b, and d are non-zero, or (4) dmay be zero while a, b, and c are non-zero.

Alternatively, any two of a, b, c, and d may be equal to zero. Forexample, (1) a and d may be zero while b and c are non-zero, or (2) band c may be zero while a and d are non-zero.

When any of the modulation scheme, error-correcting codes, and thecoding rate thereof are changed, the precoding matrix in use may also beset and changed, or the same precoding matrix may be used as-is.

Next, the baseband signal switcher 6702 from FIG. 67 is described. Thebaseband signal switcher 6702 takes weighted signal 309A and weightedsignal 316B as input, performs baseband signal switching, and outputsswitched baseband signal 6701A and switched baseband signal 6701B. Thedetails of baseband signal switching are as described with reference toFIG. 55. The baseband signal switching performed in the presentEmbodiment differs from that of FIG. 55 in terms of the signal used forswitching. The following describes the baseband signal switching of thepresent Embodiment with reference to FIG. 68.

In FIG. 68, weighted signal 309A(p1(i)) has an in-phase component I ofI_(p1)(i) and a quadrature component Q of Q_(p1)(i), while weightedsignal 316B(p2(i)) has an in-phase component I of I_(p2)(i) and aquadrature component Q of Q_(p2)(i). In contrast, switched basebandsignal 6701A(q1(i)) has an in-phase component I of I_(q1)(i) and aquadrature component Q of Q_(q1)(i), while switched baseband signal6701B(q2(i) has an in-phase component I of I_(q2)(i) and a quadraturecomponent Q of Q_(q2)(i). (Here, i represents (time or (carrier)frequency order). In the example of FIG. 67, i represents time, though imay also represent (carrier) frequency when FIG. 67 is applied to anOFDM scheme, as in FIG. 12. These points are elaborated upon below.)

Here, the baseband components are switched by the baseband signalswitcher 6702, such that:

For switched baseband signal q1(i), the in-phase component I may beI_(p1)(i) while the quadrature component Q may be Q_(p2)(i), and forswitched baseband signal q2(i), the in-phase component I may beI_(p2)(i) while the quadrature component q may be Q_(p1)(i). Themodulated signal corresponding to switched baseband signal q1(i) istransmitted by transmit antenna 1 and the modulated signal correspondingto switched baseband signal q2(i) is transmitted from transmit antenna2, simultaneously on a common frequency. As such, the modulated signalcorresponding to switched baseband signal q1(i) and the modulated signalcorresponding to switched baseband signal q2(i) are transmitted fromdifferent antennas, simultaneously on a common frequency. Alternatively,

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i) while the quadrature component may be I_(p2)(i), and forswitched baseband signal q2(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be Q_(p2)(i).

For switched baseband signal q1(i), the in-phase component may beI_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q2(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be Q_(p2)(i).

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i) while the quadrature component may be I_(p2)(i), and forswitched baseband signal q2(i), the in-phase component may be Q_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q1(i), the in-phase component may beI_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q2(i), the in-phase component may be Q_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i) while the quadrature component may be Q_(p2)(i), and forswitched baseband signal q₂(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be I_(p2)(i).

For switched baseband signal q1(i), the in-phase component may beQ_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q2(i), the in-phase component may be I_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q1(i), the in-phase component may beQ_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q2(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be I_(p2)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i) while the quadrature component may be I_(p2)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be Q_(p2)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be Q_(p2)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i) while the quadrature component may be I_(p2)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i) while the quadrature component may be Q_(p2)(i), and forswitched baseband signal q1(i), the in-phase component may be I_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i) while the quadrature component may be Q_(p2)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be I_(p2)(i).

For switched baseband signal q2(i), the in-phase component may beQ_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q1(i), the in-phase component may be I_(p2)(i)while the quadrature component may be Q_(p1)(i).

For switched baseband signal q2(i), the in-phase component may beQ_(p2)(i) while the quadrature component may be I_(p1)(i), and forswitched baseband signal q1(i), the in-phase component may be Q_(p1)(i)while the quadrature component may be I_(p2)(i).

Alternatively, the weighted signals 309A and 316B are not limited to theabove-described switching of in-phase component and quadraturecomponent. Switching may be performed on in-phase components andquadrature components greater than those of the two signals.

Also, while the above examples describe switching performed on basebandsignals having a common time (common (sub-)carrier) frequency), thebaseband signals being switched need not necessarily have a common time(common (sub-)carrier) frequency). For example, any of the following arepossible.

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w), and forswitched baseband signal q2(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be I_(p2)(i+w), and forswitched baseband signal q2(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w).

For switched baseband signal q1(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q2(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w).

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be I_(p2)(i+w), and forswitched baseband signal q2(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q1(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q2(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q1(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w), and forswitched baseband signal q2(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be I_(p2)(i+w).

For switched baseband signal q1(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q2(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q1(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q2(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be I_(p2)(i+w).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be I_(p2)(i+w), and forswitched baseband signal q1(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w).

For switched baseband signal q2(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q1(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be I_(p2)(i+w), and forswitched baseband signal q1(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q2(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q1(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w), and forswitched baseband signal q1(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q2(i), the in-phase component may beI_(p1)(i+v) while the quadrature component may be Q_(p2)(i+w), and forswitched baseband signal q1(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be I_(p2)(i+w).

For switched baseband signal q2(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q1(i), the in-phase component may beI_(p2)(i+w) while the quadrature component may be Q_(p1)(i+v).

For switched baseband signal q2(i), the in-phase component may beQ_(p2)(i+w) while the quadrature component may be I_(p1)(i+v), and forswitched baseband signal q1(i), the in-phase component may beQ_(p1)(i+v) while the quadrature component may be I_(p2)(i+w).

Here, weighted signal 309A(p1(i)) has an in-phase component I ofI_(p1)(i) and a quadrature component Q of Q_(p1)(i), while weightedsignal 316B(p2(i)) has an in-phase component I of I_(s2)(i) and aquadrature component Q of Q_(p2)(i). In contrast, switched basebandsignal 6701A(q1(i)) has an in-phase component I of I_(q1)(i) and aquadrature component Q of Q_(q1)(i), while switched baseband signal6701B(q2(i)) has an in-phase component I_(q2)(i) and a quadraturecomponent Q of Q_(q2)(i).

In FIG. 68, as described above, weighted signal 309A(p1(i)) has anin-phase component I of I_(p1)(i) and a quadrature component Q ofQ_(p1)(i), while weighted signal 316B(p2(i)) has an in-phase component Iof I_(p2)(i) and a quadrature component Q of Q_(p2)(i). In contrast,switched baseband signal 6701A(q1(i)) has an in-phase component I ofI_(q1)(i) and a quadrature component Q of Q_(q1)(i), while switchedbaseband signal 6701B(q2(i)) has an in-phase component I_(q2)(i) and aquadrature component Q of Q_(q2)(i).

As such, in-phase component I of I_(q1)(i) and quadrature component Q ofQ_(q1)(i) of switched baseband signal 6701A(q1(i)) and in-phasecomponent I_(q2)(i) and quadrature component Q of Q_(q2)(i) of basebandsignal 6701B(q2(i)) are expressible as any of the above.

As such, the modulated signal corresponding to switched baseband signal6701A(q1(i)) is transmitted from transmit antenna 312A, while themodulated signal corresponding to switched baseband signal 6701B(q2(i))is transmitted from transmit antenna 312B, both being transmittedsimultaneously on a common frequency. Thus, the modulated signalscorresponding to switched baseband signal 6701A(q1(i)) and switchedbaseband signal 6701B(q2(i)) are transmitted from different antennas,simultaneously on a common frequency.

Phase changer 317B takes switched baseband signal 6701B and signalprocessing scheme information 315 as input and regularly changes thephase of switched baseband signal 6701B for output. This regular changeis a change of phase performed according to a predetermined phasechanging pattern having a predetermined period (cycle) (e.g., every nsymbols (n being an integer, n≥1) or at a predetermined interval). Thephase changing pattern is described in detail in Embodiment 4.

Wireless unit 310B takes post-phase-change signal 309B as input andperforms processing such as quadrature modulation, band limitation,frequency conversion, amplification, and so on, then outputs transmitsignal 311B. Transmit signal 311B is then output as radio waves by anantenna 312B.

FIG. 67, much like FIG. 3, is described as having a plurality ofencoders. However, FIG. 67 may also have an encoder and a distributorlike FIG. 4. In such a case, the signals output by the distributor arethe respective input signals for the interleaver, while subsequentprocessing remains as described above for FIG. 67, despite the changesrequired thereby.

FIG. 5 illustrates an example of a frame configuration in the timedomain for a transmission device according to the present Embodiment.Symbol 500_1 is a symbol for notifying the reception device of thetransmission scheme. For example, symbol 500_1 conveys information suchas the error-correction scheme used for transmitting data symbols, thecoding rate thereof, and the modulation scheme used for transmittingdata symbols.

Symbol 5012 is for estimating channel fluctuations for modulated signalz2(t) (where t is time) transmitted by the transmission device. Symbol502_1 is a data symbol transmitted by modulated signal z1(t) as symbolnumber u (in the time domain). Symbol 503_1 is a data symbol transmittedby modulated signal z1(t) as symbol number u+1.

Symbol 501_2 is for estimating channel fluctuations for modulated signalz2(t) (where t is time) transmitted by the transmission device. Symbol502_2 is a data symbol transmitted by modulated signal z2(t) as symbolnumber u. Symbol 503_2 is a data symbol transmitted by modulated signalz1(t) as symbol number u+1.

Here, the symbols of z1(t) and of z2(t) having the same time (identicaltiming) are transmitted from the transmit antenna using the same(shared/common) frequency.

The following describes the relationships between the modulated signalsz1(t) and z2(t) transmitted by the transmission device and the receivedsignals r1(t) and r2(t) received by the reception device.

In FIG. 5, 504#1 and 504#2 indicate transmit antennas of thetransmission device, while 505#1 and 505#2 indicate receive antennas ofthe reception device. The transmission device transmits modulated signalz1(t) from transmit antenna 504#1 and transmits modulated signal z2(t)from transmit antenna 504#2. Here, modulated signals z1(t) and z2(t) areassumed to occupy the same (shared/common) frequency (bandwidth). Thechannel fluctuations in the transmit antennas of the transmission deviceand the antennas of the reception device are h₁₁(t), h₁₂(t), h₂₁(t), andh₂₂(t), respectively. Assuming that receive antenna 505#1 of thereception device receives received signal r1(t) and that receive antenna505#2 of the reception device receives received signal r2(t), thefollowing relationship holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 66} \right\rbrack & \; \\{{\begin{pmatrix}{r\; 1(t)} \\{r\; 2(t)}\end{pmatrix}\begin{pmatrix}{h_{11}(t)} & {h_{12}(t)} \\{h_{21}(t)} & {h_{22}(t)}\end{pmatrix}} = \begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}} & \left( {{formula}\mspace{14mu} 66} \right)\end{matrix}$

FIG. 69 pertains to the weighting scheme (precoding scheme), thebaseband switching scheme, and the phase changing scheme of the presentEmbodiment. The weighting unit 600 is a combined version of theweighting units 308A and 308B from FIG. 67. As shown, stream s1(t) andstream s2(t) correspond to the baseband signals 307A and 307B of FIG. 3.That is, the streams s1(t) and s2(t) are baseband signals made up of anin-phase component I and a quadrature component Q conforming to mappingby a modulation scheme such as QPSK, 16-QAM, and 64-QAM. As indicated bythe frame configuration of FIG. 69, stream s1(t) is represented as s1(u)at symbol number u, as s1(u+1) at symbol number u+1, and so forth.Similarly, stream s2(t) is represented as s2(u) at symbol number u, ass2(u+1) at symbol number u+1, and so forth. The weighting unit 600 takesthe baseband signals 307A (s1(t)) and 307B (s2(t)) as well as the signalprocessing scheme information 315 from FIG. 67 as input, performsweighting in accordance with the signal processing scheme information315, and outputs the weighted signals 309A (p₁(t)) and 316B(p₂(t)) fromFIG. 67.

Here, given vector W1=(w11,wl2) from the first row of the fixedprecoding matrix F, p₁(t) can be expressed as formula 67, below.[Math. 67]p1(t)=W1s1(t)  (formula 67)

Here, given vector W2=(w21,w22) from the first row of the fixedprecoding matrix F, p₂(t) can be expressed as formula 68, below.[Math. 68]p2(t)=W2s2(t)  (formula 68)

Accordingly, precoding matrix F may be expressed as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 69} \right\rbrack & \; \\{F = \begin{pmatrix}{w\; 11} & {w\; 12} \\{w\; 21} & {w\; 22}\end{pmatrix}} & \left( {{formula}\mspace{14mu} 69} \right)\end{matrix}$

After the baseband signals have been switched, switched baseband signal6701A(q1(i)) has an in-phase component I of Iq₁(i) and a quadraturecomponent Q of Qp₁(i), and switched baseband signal 6701B(q₂(i)) has anin-phase component I of Iq₂(i) and a quadrature component Q of Qq₂(i).The relationships between all of these are as stated above. When thephase changer uses phase changing formula y(t), the post-phase-changebaseband signal 309B(q′₂(i)) is given by formula 70, below.[Math. 70]q2′(t)=y(t)q2(t)  (formula 70)

Here, y(t) is a phase changing formula obeying a predetermined scheme.For example, given a period (cycle) of four and time u, the phasechanging formula may be expressed as formula 71, below.[Math. 71]y(u)=e ^(j0)  (formula 71)

Similarly, the phase changing formula for time u+1 may be, for example,as given by formula 72.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 72} \right\rbrack & \; \\{{y\left( {u + 1} \right)} = e^{j\frac{\pi}{2}}} & \left( {{formula}\mspace{14mu} 72} \right)\end{matrix}$

That is, the phase changing formula for time u+k generalizes to formula73.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 73} \right\rbrack & \; \\{{y\left( {u + k} \right)} = e^{j\frac{k\;\pi}{2}}} & \left( {{formula}\mspace{14mu} 73} \right)\end{matrix}$

Note that formula 71 through formula 73 are given only as an example ofa regular change of phase.

The regular change of phase is not restricted to a period (cycle) offour. Improved reception capabilities (the error-correctioncapabilities, to be exact) may potentially be promoted in the receptiondevice by increasing the period (cycle) number (this does not mean thata greater period (cycle) is better, though avoiding small numbers suchas two is likely ideal).

Furthermore, although formula 71 through formula 73, above, represent aconfiguration in which a change of phase is carried out through rotationby consecutive predetermined phases (in the above formula, every π/2),the change of phase need not be rotation by a constant amount but mayalso be random. For example, in accordance with the predetermined period(cycle) of y(t), the phase may be changed through sequentialmultiplication as shown in formula 74 and formula 75. The key point ofthe regular change of phase is that the phase of the modulated signal isregularly changed. The phase changing degree variance rate is preferablyas even as possible, such as from −π radians to π radians. However,given that this concerns a distribution, random variance is alsopossible.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 74} \right\rbrack & \; \\\left. e^{j\; 0}\rightarrow\left. e^{j\frac{\pi}{5}}\rightarrow\left. e^{j\frac{2\;\pi}{5}}\rightarrow\left. e^{j\frac{3\;\pi}{5}}\rightarrow\left. e^{j\frac{4\pi}{5}}\rightarrow\left. e^{j\;\pi}\rightarrow\left. e^{j\frac{6\pi}{5}}\rightarrow\left. e^{j\frac{7\pi}{5}}\rightarrow\left. e^{j\frac{8\pi}{5}}\rightarrow e^{j\frac{9\pi}{5}} \right. \right. \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 74} \right) \\\left\lbrack {{Math}.\mspace{14mu} 75} \right\rbrack & \; \\\left. e^{j\frac{\pi}{2}}\rightarrow\left. e^{j\;\pi}\rightarrow\left. e^{j\frac{3\;\pi}{2}}\rightarrow\left. e^{j\; 2\;\pi}\rightarrow\left. e^{j\frac{\pi}{4}}\rightarrow\left. e^{j\frac{3}{4}\pi}\rightarrow\left. e^{j\frac{5\pi}{4}}\rightarrow e^{j\frac{7\pi}{4}} \right. \right. \right. \right. \right. \right. \right. & \left( {{formula}\mspace{14mu} 75} \right)\end{matrix}$

As such, the weighting unit 600 of FIG. 6 performs precoding usingfixed, predetermined precoding weights, the baseband signal switcherperforms baseband signal switching as described above, and the phasechanger changes the phase of the signal input thereto while regularlyvarying the degree of change.

When a specialized precoding matrix is used in the LOS environment, thereception quality is likely to improve tremendously. However, dependingon the direct wave conditions, the phase and amplitude components of thedirect wave may greatly differ from the specialized precoding matrix,upon reception. The LOS environment has certain rules. Thus, datareception quality is tremendously improved through a regular change oftransmit signal phase that obeys those rules. The present inventionoffers a signal processing scheme for improving the LOS environment.

FIG. 7 illustrates a sample configuration of a reception device 700pertaining to the present embodiment. Wireless unit 703_X receives, asinput, received signal 702_X received by antenna 701_X, performsprocessing such as frequency conversion, quadrature demodulation, andthe like, and outputs baseband signal 704_X.

Channel fluctuation estimator 705_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₁₁ from formula 66, and outputs channelestimation signal 706_1.

Channel fluctuation estimator 705_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_X as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₁₂ from formula 66, and outputs channelestimation signal 706_2.

Wireless unit 703_Y receives, as input, received signal 702_Y receivedby antenna 701_X, performs processing such as frequency conversion,quadrature demodulation, and the like, and outputs baseband signal704_Y.

Channel fluctuation estimator 707_1 for modulated signal z1 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_1 for channel estimation from FIG. 5,estimates the value of h₂₁ from formula 66, and outputs channelestimation signal 708_1.

Channel fluctuation estimator 707_2 for modulated signal z2 transmittedby the transmission device takes baseband signal 704_Y as input,extracts reference symbol 501_2 for channel estimation from FIG. 5,estimates the value of h₂₂ from formula 66, and outputs channelestimation signal 708_2.

A control information decoder 709 receives baseband signal 704_X andbaseband signal 704_Y as input, detects symbol 500_1 that indicates thetransmission scheme from FIG. 5, and outputs a transmission devicetransmission scheme information signal 710.

A signal processor 711 takes the baseband signals 704_X and 704_Y, thechannel estimation signals 706_1, 706_2, 708_1, and 708_2, and thetransmission scheme information signal 710 as input, performs detectionand decoding, and then outputs received data 712_1 and 712_2.

Next, the operations of the signal processor 711 from FIG. 7 aredescribed in detail. FIG. 8 illustrates a sample configuration of thesignal processor 711 pertaining to the present embodiment. As shown, thesignal processor 711 is primarily made up of an inner MIMO detector, asoft-in/soft-out decoder, and a coefficient generator. Non-PatentLiterature 2 and Non-Patent Literature 3 describe the scheme ofiterative decoding with this structure. The MIMO system described inNon-Patent Literature 2 and Non-Patent Literature 3 is a spatialmultiplexing MIMO system, while the present Embodiment differs fromNon-Patent Literature 2 and Non-Patent Literature 3 in describing a MIMOsystem that regularly changes the phase over time, while using theprecoding matrix and performing baseband signal switching. Taking the(channel) matrix H(t) of formula 66, then by letting the precodingweight matrix from FIG. 69 be F (here, a fixed precoding matrixremaining unchanged for a given received signal) and letting the phasechanging formula used by the phase changer from FIG. 69 be Y(t) (here,Y(t) changes over time t), then given the baseband signal switching, thereceive vector R(t)=(r1(t),r2(t))^(T) and the stream vectorS(t)=(s1(t),s2(t))^(T) lead to the decoding method of Non-PatentLiterature 2 and Non-Patent Literature 3, thus enabling MIMO detection.

Accordingly, the coefficient generator 819 from FIG. 8 takes atransmission scheme information signal 818 (corresponding to 710 fromFIG. 7) indicated by the transmission device (information for specifyingthe fixed precoding matrix in use and the phase changing pattern usedwhen the phase is changed) and outputs a signal processing schemeinformation signal 820.

The inner MIMO detector 803 takes the signal processing schemeinformation signal 820 as input and performs iterative detection anddecoding using the signal. The operations are described below.

The processor illustrated in FIG. 8 uses a processing scheme, as isillustrated in FIG. 10, to perform iterative decoding (iterativedetection). First, detection of one codeword (or one frame) of modulatedsignal (stream) s1 and of one codeword (or one frame) of modulatedsignal (stream) s2 are performed. As a result, the log-likelihood ratioof each bit of the codeword (or frame) of modulated signal (stream) s1and of the codeword (or frame) of modulated signal (stream) s2 areobtained from the soft-in/soft-out decoder. Next, the log-likelihoodratio is used to perform a second round of detection and decoding. Theseoperations (referred to as iterative decoding (iterative detection)) areperformed multiple times. The following explanations center on thecreation of the log-likelihood ratio of a symbol at a specific timewithin one frame.

In FIG. 8, a memory 815 takes baseband signal 801X (corresponding tobaseband signal 704_X from FIG. 7), channel estimation signal group 802X(corresponding to channel estimation signals 706_1 and 706_2 from FIG.7), baseband signal 801Y (corresponding to baseband signal 704_Y fromFIG. 7), and channel estimation signal group 802Y (corresponding tochannel estimation signals 708_1 and 708_2 from FIG. 7) as input,performs iterative decoding (iterative detection), and stores theresulting matrix as a transformed channel signal group. The memory 815then outputs the above-described signals as needed, specifically asbaseband signal 816X, transformed channel estimation signal group 817X,baseband signal 816Y, and transformed channel estimation signal group817Y.

Subsequent operations are described separately for initial detection andfor iterative decoding (iterative detection).

(Initial Detection)

The inner MIMO detector 803 takes baseband signal 801X, channelestimation signal group 802X, baseband signal 801Y, and channelestimation signal group 802Y as input. Here, the modulation scheme formodulated signal (stream) s1 and modulated signal (stream) s2 isdescribed as 16-QAM.

The inner MIMO detector 803 first computes a candidate signal pointcorresponding to baseband signal 801X from the channel estimation signalgroups 802X and 802Y. FIG. 11 represents such a calculation. In FIG. 11,each black dot is a candidate signal point in the IQ plane. Given thatthe modulation scheme is 16-QAM, 256 candidate signal points exist.(However, FIG. 11 is only a representation and does not indicate all 256candidate signal points.) Letting the four bits transmitted in modulatedsignal s1 be b0, b1, b2, and b3 and the four bits transmitted inmodulated signal s2 be b4, b5, b6, and b7, candidate signal pointscorresponding to (b0, b1, b2, b3, b4, b5, b6, b7) are found in FIG. 11.The Euclidean squared distance between each candidate signal point andeach received signal point 1101 (corresponding to baseband signal 801X)is then computed. The Euclidian squared distance between each point isdivided by the noise variance σ². Accordingly, E_(X)(b0, b1, b2, b3, b4,b5, b6, b7) is calculated. That is, the Euclidian squared distancebetween a candidate signal point corresponding to (b0, b1, b2, b3, b4,b5, b6, b7) and a received signal point is divided by the noisevariance. Here, each of the baseband signals and the modulated signalss1 and s2 is a complex signal.

Similarly, the inner MIMO detector 803 calculates candidate signalpoints corresponding to baseband signal 801Y from channel estimationsignal group 802X and channel estimation signal group 802Y, computes theEuclidean squared distance between each of the candidate signal pointsand the received signal points (corresponding to baseband signal 801Y),and divides the Euclidean squared distance by the noise variance σ2.Accordingly, E_(Y)(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. Thatis, E_(Y) is the Euclidian squared distance between a candidate signalpoint corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a receivedsignal point, divided by the noise variance.

Next, E_(X)(b0, b1, b2, b3, b4, b5, b6, b7)+E_(Y)(b0, b1, b2, b3, b4,b5, b6, b7)=E(b0, b1, b2, b3, b4, b5, b6, b7) is computed.

The inner MIMO detector 803 outputs E(b0, b1, b2, b3, b4, b5, b6, b7) asthe signal 804.

The log-likelihood calculator 805A takes the signal 804 as input,calculates the log-likelihood of bits b0, b1, b2, and b3, and outputsthe log-likelihood signal 806A. Note that this log-likelihoodcalculation produces the log-likelihood of a bit being 1 and thelog-likelihood of a bit being 0. The calculation is as shown in formula28, formula 29, and formula 30, and the details thereof are given byNon-Patent Literature 2 and 3.

Similarly, log-likelihood calculator 805B takes the signal 804 as input,calculates the log-likelihood of bits b4, b5, b6, and b7, and outputslog-likelihood signal 806A.

A deinterleaver (807A) takes log-likelihood signal 806A as input,performs deinterleaving corresponding to that of the interleaver (theinterleaver (304A) from FIG. 67), and outputs deinterleavedlog-likelihood signal 808A.

Similarly, a deinterleaver (807B) takes log-likelihood signal 806B asinput, performs deinterleaving corresponding to that of the interleaver(the interleaver (304B) from FIG. 67), and outputs deinterleavedlog-likelihood signal 808B.

Log-likelihood ratio calculator 809A takes deinterleaved log-likelihoodsignal 808A as input, calculates the log-likelihood ratio of the bitsencoded by encoder 302A from FIG. 67, and outputs log-likelihood ratiosignal 810A.

Similarly, log-likelihood ratio calculator 809B takes deinterleavedlog-likelihood signal 808B as input, calculates the log-likelihood ratioof the bits encoded by encoder 302B from FIG. 67, and outputslog-likelihood ratio signal 810B.

Soft-in/soft-out decoder 811A takes log-likelihood ratio signal 810A asinput, performs decoding, and outputs a decoded log-likelihood ratio812A.

Similarly, soft-in/soft-out decoder 811B takes log-likelihood ratiosignal 810B as input, performs decoding, and outputs decodedlog-likelihood ratio 812B.

(Iterative Decoding (Iterative Detection), k Iterations)

The interleaver (813A) takes the k−1th decoded log-likelihood ratio 812Adecoded by the soft-in/soft-out decoder as input, performs interleaving,and outputs an interleaved log-likelihood ratio 814A. Here, theinterleaving pattern used by the interleaver (813A) is identical to thatof the interleaver (304A) from FIG. 67.

Another interleaver (813B) takes the k−1th decoded log-likelihood ratio812B decoded by the soft-in/soft-out decoder as input, performsinterleaving, and outputs interleaved log-likelihood ratio 814B. Here,the interleaving pattern used by the interleaver (813B) is identical tothat of the other interleaver (304B) from FIG. 67.

The inner MIMO detector 803 takes baseband signal 816X, transformedchannel estimation signal group 817X, baseband signal 816Y, transformedchannel estimation signal group 817Y, interleaved log-likelihood ratio814A, and interleaved log-likelihood ratio 814B as input. Here, basebandsignal 816X, transformed channel estimation signal group 817X, basebandsignal 816Y, and transformed channel estimation signal group 817Y areused instead of baseband signal 801X, channel estimation signal group802X, baseband signal 801Y, and channel estimation signal group 802Ybecause the latter cause delays due to the iterative decoding.

The iterative decoding operations of the inner MIMO detector 803 differfrom the initial detection operations thereof in that the interleavedlog-likelihood ratios 814A and 814B are used in signal processing forthe former. The inner MIMO detector 803 first calculates E(b0, b1, b2,b3, b4, b5, b6, b7) in the same manner as for initial detection. Inaddition, the coefficients corresponding to formula 11 and formula 32are computed from the interleaved log-likelihood ratios 814A and 914B.The value of E(b0, b1, b2, b3, b4, b5, b6, b7) is corrected using thecoefficients so calculated to obtain E′(b0, b1, b2, b3, b4, b5, b6, b7),which is output as the signal 804.

Log-likelihood calculator 805A takes the signal 804 as input, calculatesthe log-likelihood of bits b0, b1, b2, and b3, and outputs alog-likelihood signal 806A. Note that this log-likelihood calculationproduces the log-likelihood of a bit being 1 and the log-likelihood of abit being 0. The calculation is as shown in formula 31 through formula35, and the details are given by Non-Patent Literature 2 and 3.

Similarly, log-likelihood calculator 805B takes the signal 804 as input,calculates the log-likelihood of bits b4, b5, b6, and b7, and outputslog-likelihood signal 806B. Operations performed by the deinterleaveronwards are similar to those performed for initial detection.

While FIG. 8 illustrates the configuration of the signal processor whenperforming iterative detection, this structure is not absolutelynecessary as good reception improvements are obtainable by iterativedetection alone. As long as the components needed for iterativedetection are present, the configuration need not include theinterleavers 813A and 813B. In such a case, the inner MIMO detector 803does not perform iterative detection.

As shown in Non-Patent Literature 5 and the like, QR decomposition mayalso be used to perform initial detection and iterative detection. Also,as indicated by Non-Patent Literature 11, MMSE and ZF linear operationsmay be performed when performing initial detection.

FIG. 9 illustrates the configuration of a signal processor unlike thatof FIG. 8, that serves as the signal processor for modulated signalstransmitted by the transmission device from FIG. 4 as used in FIG. 67.The point of difference from FIG. 8 is the number of soft-in/soft-outdecoders. A soft-in/soft-out decoder 901 takes the log-likelihood ratiosignals 810A and 810B as input, performs decoding, and outputs a decodedlog-likelihood ratio 902. A distributor 903 takes the decodedlog-likelihood ratio 902 as input for distribution. Otherwise, theoperations are identical to those explained for FIG. 8.

As described above, when a transmission device according to the presentEmbodiment using a MIMO system transmits a plurality of modulatedsignals from a plurality of antennas, changing the phase over time whilemultiplying by the precoding matrix so as to regularly change the phaseresults in improvements to data reception quality for a reception devicein a LOS environment, where direct waves are dominant, compared to aconventional spatial multiplexing MIMO system.

In the present Embodiment, and particularly in the configuration of thereception device, the number of antennas is limited and explanations aregiven accordingly. However, the Embodiment may also be applied to agreater number of antennas. In other words, the number of antennas inthe reception device does not affect the operations or advantageouseffects of the present Embodiment.

Further, in the present Embodiments, the encoding is not particularlylimited to LDPC codes. Similarly, the decoding scheme is not limited toimplementation by a soft-in/soft-out decoder using sum-product decoding.The decoding scheme used by the soft-in/soft-out decoder may also be,for example, the BCJR algorithm, SOVA, and the Max-Log-Map algorithm.Details are provided in Non-Patent Literature 6.

In addition, although the present Embodiment is described using asingle-carrier scheme, no limitation is intended in this regard. Thepresent Embodiment is also applicable to multi-carrier transmission.Accordingly, the present Embodiment may also be realized using, forexample, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, waveletOFDM as described in Non-Patent Literature 7, and so on. Furthermore, inthe present Embodiment, symbols other than data symbols, such as pilotsymbols (preamble, unique word, and so on) or symbols transmittingcontrol information, may be arranged within the frame in any manner.

The following describes an example in which OFDM is used as amulti-carrier scheme.

FIG. 70 illustrates the configuration of a transmission device usingOFDM. In FIG. 70, components operating in the manner described for FIGS.3, 12, and 67 use identical reference numbers.

An OFDM-related processor 1201A takes weighted signal 309A as input,performs OFDM-related processing thereon, and outputs transmit signal1202A. Similarly, OFDM-related processor 1201B takes post-phase-changesignal 309B as input, performs OFDM-related processing thereon, andoutputs transmit signal 1202B

FIG. 13 illustrates a sample configuration of the OFDM-relatedprocessors 7001A and 1201B and onward from FIG. 70. Components 1301Athrough 1310A belong between 1201A and 312A from FIG. 70, whilecomponents 1301B through 1310B belong between 1201B and 312B.

Serial-to-parallel converter 1302A performs serial-to-parallelconversion on switched baseband signal 1301A (corresponding to switchedbaseband signal 6701A from FIG. 70) and outputs parallel signal 1303A.

Reorderer 1304A takes parallel signal 1303A as input, performsreordering thereof, and outputs reordered signal 1305A. Reordering isdescribed in detail later.

IFFT unit 1306A takes reordered signal 1305A as input, applies an IFFTthereto, and outputs post-IFFT signal 1307A.

Wireless unit 1308A takes post-IFFT signal 1307A as input, performsprocessing such as frequency conversion and amplification, thereon, andoutputs modulated signal 1309A. Modulated signal 1309A is then output asradio waves by antenna 1310A.

Serial-to-parallel converter 1302B performs serial-to-parallelconversion on post-phase-change signal 1301B (corresponding topost-phase-change signal 309B from FIG. 12) and outputs parallel signal1303B.

Reorderer 1304B takes parallel signal 1303B as input, performsreordering thereof, and outputs reordered signal 1305B. Reordering isdescribed in detail later.

IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFTthereto, and outputs post-IFFT signal 1307B.

Wireless unit 1308B takes post-IFFT signal 1307B as input, performsprocessing such as frequency conversion and amplification thereon, andoutputs modulated signal 1309B. Modulated signal 1309B is then output asradio waves by antenna 1310A.

The transmission device from FIG. 67 does not use a multi-carriertransmission scheme. Thus, as shown in FIG. 69, a change of phase isperformed to achieve a period (cycle) of four and the post-phase-changesymbols are arranged in the time domain. As shown in FIG. 70, whenmulti-carrier transmission, such as OFDM, is used, then, naturally,symbols in precoded baseband signals having undergone switching andphase changing may be arranged in the time domain as in FIG. 67, andthis may be applied to each (sub-)carrier. However, for multi-carriertransmission, the arrangement may also be in the frequency domain, or inboth the frequency domain and the time domain. The following describesthese arrangements.

FIGS. 14A and 14B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13.The frequency axes are made up of (sub-)carriers 0 through 9. Themodulated signals z1 and z2 share common time (timing) and use a commonfrequency band. FIG. 14A illustrates a reordering scheme for the symbolsof modulated signal z1, while FIG. 14B illustrates a reordering schemefor the symbols of modulated signal z2. With respect to the symbols ofswitched baseband signal 1301A input to serial-to-parallel converter1302A, the ordering is #0, #1, #2, #3, and so on. Here, given that theexample deals with a period (cycle) of four, #0, #1, #2, and #3 areequivalent to one period (cycle). Similarly, #4n, #4n+1, #4n+2, and#4n+3 (n being a non-zero positive integer) are also equivalent to oneperiod (cycle).

As shown in FIG. 14A, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement. Here, modulated signals z1 and z2 are complexsignals.

Similarly, with respect to the symbols of weighted signal 1301B input toserial-to-parallel converter 1302B, the assigned ordering is #0, #1, #2,#3, and so on. Here, given that the example deals with a period (cycle)of four, a different change in phase is applied to each of #0, #1, #2,and #3, which are equivalent to one period (cycle). Similarly, adifferent change in phase is applied to each of #4n, #4n+1, #4n+2, and#4n+3 (n being a non-zero positive integer), which are also equivalentto one period (cycle)

As shown in FIG. 14B, symbols #0, #1, #2, #3, and so on are arranged inorder, beginning at carrier 0. Symbols #0 through #9 are given time $1,followed by symbols #10 through #19 which are given time #2, and so onin a regular arrangement.

The symbol group 1402 shown in FIG. 14B corresponds to one period(cycle) of symbols when the phase changing scheme of FIG. 69 is used.Symbol #0 is the symbol obtained by using the phase at time u in FIG.69, symbol #1 is the symbol obtained by using the phase at time u+1 inFIG. 69, symbol #2 is the symbol obtained by using the phase at time u+2in FIG. 69, and symbol #3 is the symbol obtained by using the phase attime u+3 in FIG. 69. Accordingly, for any symbol #x, symbol #x is thesymbol obtained by using the phase at time u in FIG. 69 when x mod 4equals 0 (i.e., when the remainder of x divided by 4 is 0, mod being themodulo operator), symbol #x is the symbol obtained by using the phase attime x+1 in FIG. 69 when x mod 4 equals 1, symbol #x is the symbolobtained by using the phase at time x+2 in FIG. 69 when x mod 4 equals2, and symbol #x is the symbol obtained by using the phase at time x+3in FIG. 69 when x mod 4 equals 3.

In the present Embodiment, modulated signal z1 shown in FIG. 14A has notundergone a change of phase.

As such, when using a multi-carrier transmission scheme such as OFDM,and unlike single carrier transmission, symbols can be arranged in thefrequency domain. Of course, the symbol arrangement scheme is notlimited to those illustrated by FIGS. 14A and 14B. Further examples areshown in FIGS. 15A, 15B, 16A, and 16B.

FIGS. 15A and 15B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 15A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 15Billustrates a reordering scheme for the symbols of modulated signal z2.FIGS. 15A and 15B differ from FIGS. 14A and 14B in the reordering schemeapplied to the symbols of modulated signal z1 and the symbols ofmodulated signal z2. In FIG. 15B, symbols #0 through #5 are arranged atcarriers 4 through 9, symbols #6 though #9 are arranged at carriers 0through 3, and this arrangement is repeated for symbols #10 through #19.Here, as in FIG. 14B, symbol group 1502 shown in FIG. 15B corresponds toone period (cycle) of symbols when the phase changing scheme of FIG. 6is used.

FIGS. 16A and 16B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 14A and 14B. FIG. 16A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 16Billustrates a reordering scheme for the symbols of modulated signal z2.FIGS. 16A and 16B differ from FIGS. 14A and 14B in that, while FIGS. 14Aand 14B showed symbols arranged at sequential carriers, FIGS. 16A and16B do not arrange the symbols at sequential carriers. Obviously, forFIGS. 16A and 16B, different reordering schemes may be applied to thesymbols of modulated signal z1 and to the symbols of modulated signal z2as in FIGS. 15A and 15B.

FIGS. 17A and 17B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from those of FIGS. 14A through 16B. FIG. 17A illustrates areordering scheme for the symbols of modulated signal z1 while FIG. 17Billustrates a reordering scheme for the symbols of modulated signal z2.While FIGS. 14A through 16B show symbols arranged with respect to thefrequency axis, FIGS. 17A and 17B use the frequency and time axestogether in a single arrangement.

While FIG. 69 describes an example where the change of phase isperformed in a four slot period (cycle), the following example describesan eight slot period (cycle). In FIGS. 17A and 17B, the symbol group1702 is equivalent to one period (cycle) of symbols when the phasechanging scheme is used (i.e., on eight symbols) such that symbol #0 isthe symbol obtained by using the phase at time u, symbol #1 is thesymbol obtained by using the phase at time u+1, symbol #2 is the symbolobtained by using the phase at time u+2, symbol #3 is the symbolobtained by using the phase at time u+3, symbol #4 is the symbolobtained by using the phase at time u+4, symbol #5 is the symbolobtained by using the phase at time u+5, symbol #6 is the symbolobtained by using the phase at time u+6, and symbol #7 is the symbolobtained by using the phase at time u+7. Accordingly, for any symbol #x,symbol #x is the symbol obtained by using the phase at time u when x mod8 equals 0, symbol #x is the symbol obtained by using the phase at timeu+1 when x mod 8 equals 1, symbol #x is the symbol obtained by using thephase at time u+2 when x mod 8 equals 2, symbol #x is the symbolobtained by using the phase at time u+3 when x mod 8 equals 3, symbol #xis the symbol obtained by using the phase at time u+4 when x mod 8equals 4, symbol #x is the symbol obtained by using the phase at timeu+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using thephase at time u+6 when x mod 8 equals 6, and symbol #x is the symbolobtained by using the phase at time u+7 when x mod 8 equals 7. In FIGS.17A and 17B four slots along the time axis and two slots along thefrequency axis are used for a total of 4×2=8 slots, in which one period(cycle) of symbols is arranged. Here, given m×n symbols per period(cycle) (i.e., m×n different phases are available for multiplication),then n slots (carriers) in the frequency domain and m slots in the timedomain should be used to arrange the symbols of each period (cycle),such that m>n. This is because the phase of direct waves fluctuatesslowly in the time domain relative to the frequency domain. Accordingly,the present Embodiment performs a regular change of phase that reducesthe influence of steady direct waves. Thus, the phase changing period(cycle) should preferably reduce direct wave fluctuations. Accordingly,m should be greater than n. Taking the above into consideration, usingthe time and frequency domains together for reordering, as shown inFIGS. 17A and 17B, is preferable to using either of the frequency domainor the time domain alone due to the strong probability of the directwaves becoming regular. As a result, the effects of the presentinvention are more easily obtained. However, reordering in the frequencydomain may lead to diversity gain due the fact that frequency-domainfluctuations are abrupt. As such, using the frequency and time domainstogether for reordering is not always ideal.

FIGS. 18A and 18B indicate frequency on the horizontal axes and time onthe vertical axes thereof, and illustrate an example of a symbolreordering scheme used by the reorderers 1301A and 1301B from FIG. 13that differs from that of FIGS. 17A and 17B. FIG. 18A illustrates areordering scheme for the symbols of modulated signal z1, while FIG. 18Billustrates a reordering scheme for the symbols of modulated signal z2.Much like FIGS. 17A and 17B, FIGS. 18A and 18B illustrate the use of thetime and frequency axes, together. However, in contrast to FIGS. 17A and17B, where the frequency axis is prioritized and the time axis is usedfor secondary symbol arrangement, FIGS. 18A and 18B prioritize the rimeaxis and use the frequency axis for secondary symbol arrangement. InFIG. 18B, symbol group 1802 corresponds to one period (cycle) of symbolswhen the phase changing scheme is used.

In FIGS. 17A, 17B, 18A, and 18B, the reordering scheme applied to thesymbols of modulated signal z1 and the symbols of modulated signal z2may be identical or may differ as like in FIGS. 15A and 15B. Eitherapproach allows good reception quality to be obtained. Also, in FIGS.17A, 17B, 18A, and 18B, the symbols may be arranged non-sequentially asin FIGS. 16A and 16B. Either approach allows good reception quality tobe obtained.

FIG. 22 indicates frequency on the horizontal axis and time on thevertical axis thereof, and illustrates an example of a symbol reorderingscheme used by the reorderers 1301A and 1301B from FIG. 13 that differsfrom the above. FIG. 22 illustrates a regular phase changing schemeusing four slots, similar to time u through u+3 from FIG. 69. Thecharacteristic feature of FIG. 22 is that, although the symbols arereordered with respect to the frequency domain, when read along the timeaxis, a periodic shift of n (n=1 in the example of FIG. 22) symbols isapparent. The frequency-domain symbol group 2210 in FIG. 22 indicatesfour symbols to which are applied the changes of phase at time u throughu+3 from FIG. 6.

Here, symbol #0 is obtained using the change of phase at time u, symbol#1 is obtained using the change of phase at time u+1, symbol #2 isobtained using the change of phase at time u+2, and symbol #3 isobtained using the change of phase at time u+3.

Similarly, for frequency-domain symbol group 2220, symbol #4 is obtainedusing the change of phase at time u, symbol #5 is obtained using thechange of phase at time u+1, symbol #6 is obtained using the change ofphase at time u+2, and symbol #7 is obtained using the change of phaseat time u+3.

The above-described change of phase is applied to the symbol at time $1.However, in order to apply periodic shifting with respect to the timedomain, the following change of phases are applied to symbol groups2201, 2202, 2203, and 2204.

For time-domain symbol group 2201, symbol #0 is obtained using thechange of phase at time u, symbol #9 is obtained using the change ofphase at time u+1, symbol #18 is obtained using the change of phase attime u+2, and symbol #27 is obtained using the change of phase at timeu+3.

For time-domain symbol group 2202, symbol #28 is obtained using thechange of phase at time u, symbol #1 is obtained using the change ofphase at time u+1, symbol #10 is obtained using the change of phase attime u+2, and symbol #19 is obtained using the change of phase at timeu+3.

For time-domain symbol group 2203, symbol #20 is obtained using thechange of phase at time u, symbol #29 is obtained using the change ofphase at time u+1, symbol #2 is obtained using the change of phase attime u+2, and symbol #11 is obtained using the change of phase at timeu+3.

For time-domain symbol group 2204, symbol #12 is obtained using thechange of phase at time u, symbol #21 is obtained using the change ofphase at time u+1, symbol #30 is obtained using the change of phase attime u+2, and symbol #3 is obtained using the change of phase at timeu+3.

The characteristic feature of FIG. 22 is seen in that, taking symbol #11as an example, the two neighbouring symbols thereof along the frequencyaxis (#10 and #12) are both symbols change using a different phase thansymbol #11, and the two neighbouring symbols thereof having the samecarrier in the time domain (#2 and #20) are both symbols changed using adifferent phase than symbol #11. This holds not only for symbol #11, butalso for any symbol having two neighboring symbols in the frequencydomain and the time domain. Accordingly, the change of phase iseffectively carried out. This is highly likely to improve data receptionquality as influence from regularizing direct waves is less prone toreception.

Although FIG. 22 illustrates an example in which n=1, the invention isnot limited in this manner. The same may be applied to a case in whichn=3. Furthermore, although FIG. 22 illustrates the realization of theabove-described effects by arranging the symbols in the frequency domainand advancing in the time domain so as to achieve the characteristiceffect of imparting a periodic shift to the symbol arrangement order,the symbols may also be randomly (or regularly) arranged to the sameeffect.

Although the present Embodiment describes a variation of Embodiment 1 inwhich a baseband signal switcher is inserted before the change of phase,the present Embodiment may also be realized as a combination withEmbodiment 2, such that the baseband signal switcher is inserted beforethe change of phase in FIGS. 26 and 28. Accordingly, in FIG. 26, phasechanger 317A takes switched baseband signal 6701A(q₁(i)) as input, andphase changer 317B takes switched baseband signal 6701B(q₂(i)) as input.The same applies to the phase changers 317A and 317B from FIG. 28.

The following describes a scheme for allowing the reception device toobtain good received signal quality for data, regardless of thereception device arrangement, by considering the location of thereception device with respect to the transmission device.

FIG. 31 illustrates an example of frame configuration for a portion ofthe symbols within a signal in the time-frequency domains, given atransmission scheme where a regular change of phase is performed for amulti-carrier scheme such as OFDM.

FIG. 31 illustrates the frame configuration of modulated signal z2′corresponding to the switched baseband signal input to phase changer317B from FIG. 67. Each square represents one symbol (although bothsignals s1 and s2 are included for precoding purposes, depending on theprecoding matrix, only one of signals s1 and s2 may be used).

Consider symbol 3100 at carrier 2 and time $2 of FIG. 31. The carrierhere described may alternatively be termed a sub-carrier.

Within carrier 2, there is a very strong correlation between the channelconditions for symbol 610A at carrier 2, time $2 and the channelconditions for the time domain nearest-neighbour symbols to time $2,i.e., symbol 3013 at time $1 and symbol 3101 at time $3 within carrier2.

Similarly, for time $2, there is a very strong correlation between thechannel conditions for symbol 3100 at carrier 2, time $2 and the channelconditions for the frequency-domain nearest-neighbour symbols to carrier2, i.e., symbol 3104 at carrier 1, time $2 and symbol 3104 at time $2,carrier 3.

As described above, there is a very strong correlation between thechannel conditions for symbol 3100 and the channel conditions for eachsymbol 3101, 3102, 3103, and 3104.

The present description considers N different phases (N being aninteger, N≥2) for multiplication in a transmission scheme where thephase is regularly changed. The symbols illustrated in FIG. 31 areindicated as e^(j0), for example. This signifies that this symbol issignal z2′ from FIG. 6 having undergone a change in phase throughmultiplication by e^(j0). That is, the values given for the symbols inFIG. 31 are the value of y(t) as given by formula 70.

The present Embodiment takes advantage of the high correlation inchannel conditions existing between neighbouring symbols in thefrequency domain and/or neighbouring symbols in the time domain in asymbol arrangement enabling high data reception quality to be obtainedby the reception device receiving the post-phase-change symbols.

In order to achieve this high data reception quality, conditions #D1-1and #D1-2 should preferably be met.

(Condition #D1-1)

As shown in FIG. 69, for a transmission scheme involving a regularchange of phase performed on switched baseband signal q2 using amulti-carrier scheme such as OFDM, time X, carrier Y is a symbol fortransmitting data (hereinafter, data symbol), neighbouring symbols inthe time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Yare also data symbols, and a different change of phase should beperformed on switched baseband signal q2 corresponding to each of thesethree data symbols, i.e., on switched baseband signal q2 at time X,carrier Y, at time X−1, carrier Y and at time X+1, carrier Y.

(Condition #D1-2)

As shown in FIG. 69, for a transmission scheme involving a regularchange of phase performed on switched baseband signal q2 using amulti-carrier scheme such as OFDM, time X, carrier Y is a symbol fortransmitting data (hereinafter, data symbol), neighbouring symbols inthe time domain, i.e., at time X, carrier Y+1 and at time X, carrier Y−1are also data symbols, and a different change of phase should beperformed on switched baseband signal q2 corresponding to each of thesethree data symbols, i.e., on switched baseband signal q2 at time X,carrier Y, at time X, carrier Y−1 and at time X, carrier Y+1.

Ideally, a data symbol should satisfy Condition #D1-1. Similarly, thedata symbols should satisfy Condition #D1-2.

The reasons supporting Conditions #D1-1 and #D1-2 are as follows.

A very strong correlation exists between the channel conditions of givensymbol of a transmit signal (hereinafter, symbol A) and the channelconditions of the symbols neighbouring symbol A in the time domain, asdescribed above.

Accordingly, when three neighbouring symbols in the time domain eachhave different phases, then despite reception quality degradation in theLOS environment (poor signal quality caused by degradation in conditionsdue to phase relations despite high signal quality in terms of SNR) forsymbol A, the two remaining symbols neighbouring symbol A are highlylikely to provide good reception quality. As a result, good receivedsignal quality is achievable after error correction and decoding.

Similarly, a very strong correlation exists between the channelconditions of given symbol of a transmit signal (symbol A) and thechannel conditions of the symbols neighbouring symbol A in the frequencydomain, as described above.

Accordingly, when three neighbouring symbols in the frequency domaineach have different phases, then despite reception quality degradationin the LOS environment (poor signal quality caused by degradation inconditions due to direct wave phase relationships despite high signalquality in terms of SNR) for symbol A, the two remaining symbolsneighbouring symbol A are highly likely to provide good receptionquality. As a result, good received signal quality is achievable aftererror correction and decoding.

Combining Conditions #D1-1 and #D1-2, ever greater data receptionquality is likely achievable for the reception device. Accordingly, thefollowing Condition #D1-3 can be derived.

(Condition #D1-3)

As shown in FIG. 69, for a transmission scheme involving a regularchange of phase performed on switched baseband signal q2 using amulti-carrier scheme such as OFDM, time X, carrier Y is a symbol fortransmitting data (data symbol), neighbouring symbols in the timedomain, i.e., at time X−1, carrier Y and at time X+1, carrier Y are alsodata symbols, and neighbouring symbols in the frequency domain, i.e., attime X, carrier Y−1 and at time X, carrier Y+1 are also data symbols,such that a different change of phase should be performed on switchedbaseband signal q2 corresponding to each of these five data symbols,i.e., on switched baseband signal q2 at time X, carrier Y, at time X,carrier Y−1, at time X, carrier Y+1, at time X−1, carrier Y and at timeX+1, carrier Y.

Here, the different changes in phase are as follows. Phase changes aredefined from 0 radians to 2π radians. For example, for time X, carrierY, a phase change of e^(jθX,Y) is applied to precoded baseband signal q₂from FIG. 69, for time X−1, carrier Y, a phase change of e^(jθX−1,Y) isapplied to precoded baseband signal q2 from FIG. 69, for time X+1,carrier Y, a phase change of e^(jθX+1,Y) is applied to precoded basebandsignal q2 from FIG. 69, such that 0≤θ_(X,Y)<2π, 0≤θ_(X−1,Y)<2π, and0≤θ_(X+1,Y)<2π, □□ all units being in radians. And, for Condition #D1-1,it follows that θ_(X,Y)≠θ_(X−1,Y), θ_(X,Y)≠θ_(X+1,Y), and thatθ_(X−1,Y)≠θ_(X+1,Y). Similarly, for Condition #D1-2, it follows thatθ_(X,Y)≠θ_(X,Y−1), θ_(X,Y)≠θ_(X,Y+1), and that θ_(X,Y−1)≠θ_(X,Y+1). And,for Condition #D1-3, it follows that θ_(X,Y)≠θ_(X−1,Y),θ_(X,Y)≠θ_(X+1,Y), θ_(X,Y)≠θ_(X,Y−1), θ_(X,Y)≠θ_(X,Y+1),θ_(X−1,Y)≠θ_(X+1,Y), θ_(X−1,Y)≠θ_(X,Y−1), θ_(X−1,Y)≠θ_(X,Y+1),θ_(X+1,Y)≠θ_(X,Y−1), θ_(X+1,Y)≠θ_(X,Y+1), and that θ_(X,Y−1)≠θ_(X,Y+1).

Ideally, a data symbol should satisfy Condition #D1-1.

FIG. 31 illustrates an example of Condition #D1-3, where symbol Acorresponds to symbol 3100. The symbols are arranged such that the phaseby which switched baseband signal q2 from FIG. 69 is multiplied differsfor symbol 3100, for both neighbouring symbols thereof in the timedomain 3101 and 3102, and for both neighbouring symbols thereof in thefrequency domain 3102 and 3104. Accordingly, despite received signalquality degradation of symbol 3100 for the receiver, good signal qualityis highly likely for the neighbouring signals, thus guaranteeing goodsignal quality after error correction.

FIG. 32 illustrates a symbol arrangement obtained through phase changesunder these conditions.

As evident from FIG. 32, with respect to any data symbol, a differentchange in phase is applied to each neighbouring symbol in the timedomain and in the frequency domain. As such, the ability of thereception device to correct errors may be improved.

In other words, in FIG. 32, when all neighbouring symbols in the timedomain are data symbols, Condition #D1-1 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols, Condition #D1-2 is satisfied for all Xs and allYs.

Similarly, in FIG. 32, when all neighbouring symbols in the frequencydomain are data symbols and all neighbouring symbols in the time domainare data symbols, Condition #D1-3 is satisfied for all Xs and all Ys.

The following discusses the above-described example for a case where thechange of phase is performed on two switched baseband signals q1 and q2(see FIG. 68).

Several phase changing schemes are applicable to performing a change ofphase on two switched baseband signals q1 and q2. The details thereofare explained below.

Scheme 1 involves a change of phase of switched baseband signal q2 asdescribed above, to achieve the change of phase illustrated by FIG. 32.In FIG. 32, a change of phase having a period (cycle) of ten is appliedto switched baseband signal q2. However, as described above, in order tosatisfy Conditions #D1-1, #D1-2, and #D1-3, the change in phase appliedto switched baseband signal q2 at each (sub-)carrier changes over time.(Although such changes are applied in FIG. 32 with a period (cycle) often, other phase changing schemes are also applicable.) Then, as shownin FIG. 33, the phase change degree performed on switched basebandsignal q2 produce a constant value that is one-tenth that of the changein phase performed on switched baseband signal q2. In FIG. 33, for aperiod (cycle) (of phase change performed on switched baseband signalq2) including time $1, the value of the change in phase performed onswitched baseband signal q1 is e^(j0). Then, for the next period (cycle)(of change in phase performed on switched baseband signal q2) includingtime $2, the value of the phase changing degree performed on precodedbaseband signal q1 is e^(jπ/9), and so on.

The symbols illustrated in FIG. 33 are indicated as e^(j0), for example.This signifies that this symbol is signal q1 from FIG. 26 havingundergone a change of phase through multiplication by e^(j0).

As shown in FIG. 33, the change in phase applied to switched basebandsignal q1 produces a constant value that is one-tenth that of the changein phase performed on precoded, switched baseband signal q2 such thatthe phase changing value varies with the number of each period (cycle).(As described above, in FIG. 33, the value is e^(j0) for the firstperiod (cycle), e^(jπ/9) for the second period (cycle), and so on.)

As described above, the change in phase performed on switched basebandsignal q2 has a period (cycle) of ten, but the period (cycle) can beeffectively made greater than ten by taking the degree of phase changeapplied to switched baseband signal q1 and to switched baseband signalq2 into consideration. Accordingly, data reception quality may beimproved for the reception device.

Scheme 2 involves a change in phase of switched baseband signal q2 asdescribed above, to achieve the change in phase illustrated by FIG. 32.In FIG. 32, a change of phase having a period (cycle) of ten is appliedto switched baseband signal q2. However, as described above, in order tosatisfy Conditions #D1-1, #D1-2, and #D1-3, the change in phase appliedto switched baseband signal q2 at each (sub-)carrier changes over time.(Although such changes are applied in FIG. 32 with a period (cycle) often, other phase changing schemes are also applicable.) Then, as shownin FIG. 33, the change in phase performed on switched baseband signal q2produces a constant value that is one-tenth of that performed onswitched baseband signal q2.

The symbols illustrated in FIG. 30 are indicated as e^(j0), for example.This signifies that this symbol is switched baseband signal q1 havingundergone a change of phase through multiplication by e^(j0).

As described above, the change in phase performed on switched basebandsignal q₂ has a period (cycle) of ten, but the period (cycle) can beeffectively made greater than ten by taking the changes in phase appliedto switched baseband signal q1 and to switched baseband signal q2 intoconsideration. Accordingly, data reception quality may be improved forthe reception device. An effective way of applying scheme 2 is toperform a change in phase on switched baseband signal q1 with a period(cycle) of N and perform a change in phase on precoded baseband signalq2 with a period (cycle) of M such that N and M are coprime. As such, bytaking both switched baseband signals q1 and q2 into consideration, aperiod (cycle) of N×M is easily achievable, effectively making theperiod (cycle) greater when N and M are coprime.

While the above discusses an example of the above-described phasechanging scheme, the present invention is not limited in this manner.The change in phase may be performed with respect to the frequencydomain, the time domain, or on time-frequency blocks. Similarimprovement to the data reception quality can be obtained for thereception device in all cases.

The same also applies to frames having a configuration other than thatdescribed above, where pilot symbols (SP symbols) and symbolstransmitting control information are inserted among the data symbols.The details of the change in phase in such circumstances are as follows.

FIGS. 47A and 47B illustrate the frame configuration of modulatedsignals (switched baseband signals q1 and q2) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 47A illustrates the frame configuration ofmodulated signal (switched baseband signal q1) z1 or z1′ while FIG. 47Billustrates the frame configuration of modulated signal (switchedbaseband signal q2) z2′. In FIGS. 47A and 47B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich switching or switching and change in phase have been performed.

FIGS. 47A and 47B, like FIG. 69, indicate the arrangement of symbolswhen a change in phase is applied to switched baseband signal q2 (whileno change in phase is performed on switched baseband signal q1).(Although FIG. 69 illustrates a change in phase with respect to the timedomain, switching time t with carrier f in FIG. 69 corresponds to achange in phase with respect to the frequency domain. In other words,replacing (t) with (t, f) where t is time and f is frequency correspondsto performing a change of phase on time-frequency blocks.) Accordingly,the numerical values indicated in FIGS. 47A and 47B for each of thesymbols are the values of switched baseband signal q2 after the changein phase. No values are given for the symbols of switched basebandsignal q1 (z1) from FIGS. 47A and 47B as no change in phase is performedthereon.

The important point of FIGS. 47A and 47B is that the change in phaseperformed on the data symbols of switched baseband signal q2, i.e., onsymbols having undergone precoding or precoding and switching. (Thesymbols under discussion, being precoded, actually include both symbolss1 and s2.) Accordingly, no change in phase is performed on the pilotsymbols inserted in z2′.

FIGS. 48A and 48B illustrate the frame configuration of modulatedsignals (switched baseband signals q1 and q2) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 48A illustrates the frame configuration ofmodulated signal (switched baseband signal q1) z1 or z1′ while FIG. 48Billustrates the frame configuration of modulated signal (switchedbaseband signal q2) z2′. In FIGS. 48A and 48B, 4701 marks pilot symbolswhile 4702 marks data symbols. The data symbols 4702 are symbols onwhich precoding or precoding and a change in phase have been performed.

FIGS. 48A and 48B indicate the arrangement of symbols when a change inphase is applied to switched baseband signal q1 and to switched basebandsignal q2. Accordingly, the numerical values indicated in FIGS. 48A and48B for each of the symbols are the values of switched baseband signalsq1 and q2 after the change in phase.

The important point of FIGS. 48A and 48B is that the change in phase isperformed on the data symbols of switched baseband signal q1, that is,on the precoded or precoded and switched symbols thereof, and on thedata symbols of switched baseband signal q2, that is, on the precoded orprecoded and switched symbols thereof. (The symbols under discussion,being precoded, actually include both symbols s1 and s2.) Accordingly,no change in phase is performed on the pilot symbols inserted in z1′,nor on the pilot symbols inserted in z2′.

FIGS. 49A and 49B illustrate the frame configuration of modulatedsignals (switched baseband signals q1 and q2) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 49A illustrates the frame configuration ofmodulated signal (switched baseband signal q1) z1 or z1′ while FIG. 49Billustrates the frame configuration of modulated signal (switchedbaseband signal q2) z2′. In FIGS. 49A and 49B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on which precodingor precoding and a change in phase have been performed. FIGS. 49A and49B differ from FIGS. 47A and 47B in the configuration scheme forsymbols other than data symbols. The times and carriers at which pilotsymbols are inserted into modulated signal z1′ are null symbols inmodulated signal z2′. Conversely, the times and carriers at which pilotsymbols are inserted into modulated signal z2′ are null symbols inmodulated signal z1′.

FIGS. 49A and 49B, like FIG. 69, indicate the arrangement of symbolswhen a change in phase is applied to switched baseband signal q2 (whileno change in phase is performed on switched baseband signal q1).(Although FIG. 69 illustrates a change in phase with respect to the timedomain, switching time t with carrier f in FIG. 6 corresponds to achange in phase with respect to the frequency domain. In other words,replacing (t) with (t, f) where t is time and f is frequency correspondsto performing the change of phase on time-frequency blocks.)Accordingly, the numerical values indicated in FIGS. 49A and 49B foreach of the symbols are the values of switched baseband signal q₂ afterthe change in phase. No values are given for the symbols of switchedbaseband signal q1 from FIGS. 49A and 49B as no change in phase isperformed thereon.

The important point of FIGS. 49A and 49B is that the change in phaseperformed on the data symbols of switched baseband signal q2, i.e., onsymbols having undergone precoding or precoding and switching. (Thesymbols under discussion, being precoded, actually include both symbolss1 and s2.) Accordingly, no change in phase is performed on the pilotsymbols inserted in z2′.

FIGS. 50A and 50B illustrate the frame configuration of modulatedsignals (switched baseband signals q1 and q2) z1 or z1′ and z2′ in thetime-frequency domain. FIG. 50A illustrates the frame configuration ofmodulated signal (switched baseband signal q1) z1 or z1′ while FIG. 50Billustrates the frame configuration of modulated signal (switchedbaseband signal q2) z2′. In FIGS. 50A and 50B, 4701 marks pilot symbols,4702 marks data symbols, and 4901 marks null symbols for which thein-phase component of the baseband signal I=0 and the quadraturecomponent Q=0. As such, data symbols 4702 are symbols on which precodingor precoding and a change in phase have been performed. FIGS. 50A and50B differ from FIGS. 48A and 48B in the configuration scheme forsymbols other than data symbols. The times and carriers at which pilotsymbols are inserted into modulated signal z1′ are null symbols inmodulated signal z2′. Conversely, the times and carriers at which pilotsymbols are inserted into modulated signal z2′ are null symbols inmodulated signal z1′.

FIGS. 50A and 50B indicate the arrangement of symbols when a change inphase is applied to switched baseband signal q1 and to switched basebandsignal q2. Accordingly, the numerical values indicated in FIGS. 50A and50B for each of the symbols are the values of switched baseband signalsq1 and q2 after a change in phase.

The important point of FIGS. 50A and 50B is that a change in phase isperformed on the data symbols of switched baseband signal q1, that is,on the precoded or precoded and switched symbols thereof, and on thedata symbols of switched baseband signal q2, that is, on the precoded orprecoded and switched symbols thereof. (The symbols under discussion,being precoded, actually include both symbols s1 and s2.) Accordingly,no change in phase is performed on the pilot symbols inserted in z1′,nor on the pilot symbols inserted in z2′.

FIG. 51 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 47A, 47B, 49A, and 49B. Components thereofperforming the same operations as those of FIG. 4 use the same referencesymbols thereas. FIG. 51 does not include a baseband signal switcher asillustrated in FIGS. 67 and 70. However, FIG. 51 may also include abaseband signal switcher between the weighting units and phase changers,much like FIGS. 67 and 70.

In FIG. 51, the weighting units 308A and 308B, phase changer 317B, andbaseband signal switcher only operate at times indicated by the frameconfiguration signal 313 as corresponding to data symbols.

In FIG. 51, a pilot symbol generator 5101 (that also generates nullsymbols) outputs baseband signals 5102A and 5102B for a pilot symbolwhenever the frame configuration signal 313 indicates a pilot symbol(and a null symbol).

Although not indicated in the frame configurations from FIGS. 47Athrough 50B, when precoding (and phase rotation) is not performed, suchas when transmitting a modulated signal using only one antenna (suchthat the other antenna transmits no signal) or when using a space-timecoding transmission scheme (particularly, space-time block coding) totransmit control information symbols, then the frame configurationsignal 313 takes control information symbols 5104 and controlinformation 5103 as input. When the frame configuration signal 313indicates a control information symbol, baseband signals 5102A and 5102Bthereof are output.

The wireless units 310A and 310B of FIG. 51 take a plurality of basebandsignals as input and select a desired baseband signal according to theframe configuration signal 313. The wireless units 310A and 310B thenapply OFDM signal processing and output modulated signals 311A and 311Bconforming to the frame configuration.

FIG. 52 illustrates a sample configuration of a transmission devicegenerating and transmitting modulated signal having the frameconfiguration of FIGS. 48A, 48B, 50A, and 50B. Components thereofperforming the same operations as those of FIGS. 4 and 51 use the samereference symbols thereas. FIG. 52 features an additional phase changer317A that only operates when the frame configuration signal 313indicates a data symbol. At all other times, the operations areidentical to those explained for FIG. 51. FIG. 52 does not include abaseband signal switcher as illustrated in FIGS. 67 and 70. However,FIG. 52 may also include a baseband signal switcher between theweighting unit and phase changer, much like FIGS. 67 and 70.

FIG. 53 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 51. FIG. 53 does not include a baseband signalswitcher as illustrated in FIGS. 67 and 70. However, FIG. 53 may alsoinclude a baseband signal switcher between the weighting unit and phasechanger, much like FIGS. 67 and 70. The following describes the pointsof difference. As shown in FIG. 53, phase changer 317B takes a pluralityof baseband signals as input. Then, when the frame configuration signal313 indicates a data symbol, phase changer 317B performs the change inphase on precoded baseband signal 316B. When frame configuration signal313 indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

A selector 5301 takes the plurality of baseband signals as input andselects a baseband signal having a symbol indicated by the frameconfiguration signal 313 for output.

FIG. 54 illustrates a sample configuration of a transmission device thatdiffers from that of FIG. 52. FIG. 54 does not include a baseband signalswitcher as illustrated in FIGS. 67 and 70. However, FIG. 54 may alsoinclude a baseband signal switcher between the weighting unit and phasechanger, much like FIGS. 67 and 70. The following describes the pointsof difference. As shown in FIG. 54, phase changer 317B takes a pluralityof baseband signals as input. Then, when the frame configuration signal313 indicates a data symbol, phase changer 317B performs the change inphase on precoded baseband signal 316B. When frame configuration signal313 indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 317B pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

Similarly, as shown in FIG. 54, phase changer 5201 takes a plurality ofbaseband signals as input. Then, when the frame configuration signal 313indicates a data symbol, phase changer 5201 performs the change in phaseon precoded baseband signal 309A. When frame configuration signal 313indicates a pilot symbol (or null symbol) or a control informationsymbol, phase changer 5201 pauses phase changing operations such thatthe symbols of the baseband signal are output as-is. (This may beinterpreted as performing forced rotation corresponding to e^(j0).)

The above explanations are given using pilot symbols, control symbols,and data symbols as examples. However, the present invention is notlimited in this manner. When symbols are transmitted using schemes otherthan precoding, such as single-antenna transmission or transmissionusing space-time block codes, the absence of change in phase isimportant. Conversely, performing the change of phase on symbols thathave been precoded is the key point of the present invention.

Accordingly, a characteristic feature of the present invention is thatthe change in phase is not performed on all symbols within the frameconfiguration in the time-frequency domain, but only performed onbaseband signals that have been precoded and have undergone switching.

The following describes a scheme for regularly changing the phase whenencoding is performed using block codes as described in Non-PatentLiterature 12 through 15, such as QC LDPC Codes (not only QC-LDPC butalso LDPC codes may be used), concatenated LDPC and BCH codes, Turbocodes or Duo-Binary Turbo Codes using tail-biting, and so on. Thefollowing example considers a case where two streams s1 and s2 aretransmitted. When encoding has been performed using block codes andcontrol information and the like is not necessary, the number of bitsmaking up each coded block matches the number of bits making up eachblock code (control information and so on described below may yet beincluded). When encoding has been performed using block codes or thelike and control information or the like (e.g., CRC transmissionparameters) is necessary, then the number of bits making up each codedblock is the sum of the number of bits making up the block codes and thenumber of bits making up the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed intwo coded blocks when block codes are used. Unlike FIGS. 69 and 70, forexample, FIG. 34 illustrates the varying numbers of symbols and slotsneeded in each coded block when block codes are used when, for example,two streams s1 and s2 are transmitted as indicated in FIG. 4, with anencoder and distributor. (Here, the transmission scheme may be anysingle-carrier scheme or multi-carrier scheme such as OFDM.) As shown inFIG. 34, when block codes are used, there are 6000 bits making up asingle coded block. In order to transmit these 6000 bits, the number ofrequired symbols depends on the modulation scheme, being 3000 for QPSK,1500 for 16-QAM, and 1000 for 64-QAM.

Then, given that the above-described transmission device transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols (hereinafter, slots) are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up one coded block, andwhen the modulation scheme is 64-QAM, 500 slots are needed to transmitall of the bits making up one coded block.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase. Here, five different phase changing values (or phasechanging sets) are assumed as having been prepared for use in the schemefor a regular change of phase. That is, the phase changer of theabove-described transmission device uses five phase changing values (orphase changing sets) to achieve the period (cycle) of five. (As in FIG.69, five phase changing values are needed in order to perform a changeof phase having a period (cycle) of five on switched baseband signal q2only. Similarly, in order to perform the change in phase on bothswitched baseband signals q1 and q2, two phase changing values areneeded for each slot. These two phase changing values are termed a phasechanging set. Accordingly, here, in order to perform a change of phasehaving a period (cycle) of five, five such phase changing sets should beprepared). The five phase changing values (or phase changing sets) areexpressed as PHASE[0], PHASE[1], PHASE[2], PHASE[3], and PHASE[4].

For the above-described 1500 slots needed to transmit the 6000 bitsmaking up a single coded block when the modulation scheme is QPSK,PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2]is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] isused on 300 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Furthermore, for the above-described 750 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is16-QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 150 slots,PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, andPHASE[4] is used on 150 slots.

Further still, for the above-described 500 slots needed to transmit the6000 bits making up a single coded block when the modulation scheme is64-QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 100 slots,PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, andPHASE[4] is used on 100 slots.

As described above, a scheme for a regular change of phase requires thepreparation of N phase changing values (or phase changing sets) (wherethe N different phases are expressed as PHASE[0], PHASE[1], PHASE[2] . .. PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bitsmaking up a single coded block, PHASE[0] is used on K₀ slots, PHASE[1]is used on K₁ slots, PHASE[i] is used on K_(i) slots (where i=0, 1, 2 .. . N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1]is used on K_(N−1) slots, such that Condition #D1-4 is met.

(Condition #D1-4)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (for ∀a and ∀bwhere a, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #D1-4 ispreferably satisfied for the supported modulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #D1-4 may not be satisfied for some modulation schemes. Insuch a case, the following condition applies instead of Condition #D1-4.

(Condition #D1-5)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

FIG. 35 illustrates the varying numbers of symbols and slots needed intwo coded block when block codes are used. FIG. 35 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the transmission device from FIG. 67 andFIG. 70, and the transmission device has two encoders. (Here, thetransmission scheme may be any single-carrier scheme or multi-carrierscheme such as OFDM.)

As shown in FIG. 35, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM.

The transmission device from FIG. 67 and the transmission device fromFIG. 70 each transmit two streams at once, and have two encoders. Assuch, the two streams each transmit different code blocks. Accordingly,when the modulation scheme is QPSK, two coded blocks drawn from s1 ands2 are transmitted within the same interval, e.g., a first coded blockdrawn from s1 is transmitted, then a second coded block drawn from s2 istransmitted. As such, 3000 slots are needed in order to transmit thefirst and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 slotsare needed to transmit all of the bits making up the two coded blocks,and when the modulation scheme is 64-QAM, 1000 slots are needed totransmit all of the bits making up the two coded blocks.

The following describes the relationship between the above-defined slotsand the phase of multiplication, as pertains to schemes for a regularchange of phase.

Here, five different phase changing values (or phase changing sets) areassumed as having been prepared for use in the scheme for a regularchange of phase. That is, the phase changer of the transmission devicefrom FIG. 67 and FIG. 67 uses five phase changing values (or phasechanging sets) to achieve the period (cycle) of five. (As in FIG. 69,five phase changing values are needed in order to perform a change ofphase having a period (cycle) of five on switched baseband signal q2only. Similarly, in order to perform the change in phase on bothswitched baseband signals q1 and q2, two phase changing values areneeded for each slot. These two phase changing values are termed a phasechanging set. Accordingly, here, in order to perform a change of phasehaving a period (cycle) of five, five such phase changing sets should beprepared). The five phase changing values (or phase changing sets) areexpressed as PHASE[0], PHASE[1], PHASE[2], PHASE[3], and PHASE[4].

For the above-described 3000 slots needed to transmit the 6000×2 bitsmaking up the two coded blocks when the modulation scheme is QPSK,PHASE[0] is used on 600 slots, PHASE[1] is used on 600 slots, PHASE[2]is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] isused on 600 slots. This is due to the fact that any bias in phase usagecauses great influence to be exerted by the more frequently used phase,and that the reception device is dependent on such influence for datareception quality.

Further, in order to transmit the first coded block, PHASE[0] is used onslots 600 times, PHASE[1] is used on slots 600 times, PHASE[2] is usedon slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] isused on slots 600 times. Furthermore, in order to transmit the secondcoded block, PHASE[0] is used on slots 600 times, PHASE[1] is used onslots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is usedon slots 600 times, and PHASE[4] is used on slots 600 times.

Similarly, for the above-described 1500 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is16-QAM, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots,PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, andPHASE[4] is used on 300 slots.

Further, in order to transmit the first coded block, PHASE[0] is used onslots 300 times, PHASE[1] is used on slots 300 times, PHASE[2] is usedon slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] isused on slots 300 times. Furthermore, in order to transmit the secondcoded block, PHASE[0] is used on slots 300 times, PHASE[1] is used onslots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is usedon slots 300 times, and PHASE[4] is used on slots 300 times.

Similarly, for the above-described 1000 slots needed to transmit the6000×2 bits making up the two coded blocks when the modulation scheme is64-QAM, PHASE[0] is used on 200 slots, PHASE[1] is used on 200 slots,PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, andPHASE[4] is used on 200 slots.

Further, in order to transmit the first coded block, PHASE[0] is used onslots 200 times, PHASE[1] is used on slots 200 times, PHASE[2] is usedon slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] isused on slots 200 times. Furthermore, in order to transmit the secondcoded block, PHASE[0] is used on slots 200 times, PHASE[1] is used onslots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is usedon slots 200 times, and PHASE[4] is used on slots 200 times.

As described above, a scheme for a regular change of phase requires thepreparation of N phase changing values (or phase changing sets) (wherethe N different phases are expressed as PHASE[0], PHASE[1], PHASE[2] . .. PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bitsmaking up a single coded block, PHASE[0] is used on K₀ slots, PHASE[1]is used on K₁ slots, PHASE[i] is used on K_(i) slots (where i=0, 1, 2 .. . N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1]is used on K_(N−1) slots, such that Condition #D1-6 is met.

(Condition #D1-6)

K₀=K₁ . . . =K_(i)= . . . K_(N−1). That is, K_(a)=K_(b) (for ∀a and ∀bwhere a, b, =0, 1, 2 . . . N−1 (a denotes an integer that satisfies0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).

Further, in order to transmit all of the bits making up the first codedblock, PHASE[0] is used K_(0,1) times, PHASE[1] is used K_(1,1) times,PHASE[i] is used K_(i,1) times (where i=0, 1, 2 . . . N−1 (i denotes aninteger that satisfies 0≤i≤N−1)), and PHASE[N−1] is used K_(N−1,1)times, such that Condition #D1-7 is met.

(Condition #D1-7)

K_(0,1)=K_(1,1)= . . . K_(i,1)= . . . K_(N−1,1). That is,K_(a,1)=K_(b,1) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

Furthermore, in order to transmit all of the bits making up the secondcoded block, PHASE[0] is used K_(0,2) times, PHASE[1] is used K_(1,2)times, PHASE[i] is used K_(i,2) times (where i=0, 1, 2 . . . N−1 (idenotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is usedK_(N−1,2) times, such that Condition #D1-8 is met.

(Condition #D1-8)

K_(0,2)=K_(1,2)= . . . K_(i,2)= . . . K_(N−1,2). That is,K_(a,2)=K_(b,2) (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a denotes aninteger that satisfies 0≤a≤N−1, b denotes an integer that satisfies0≤b≤N−1), a≠b).

Then, when a communication system that supports multiple modulationschemes selects one such supported scheme for use, Condition #D1-6Condition #D1-7, and Condition #D1-8 are preferably satisfied for thesupported modulation scheme.

However, when multiple modulation schemes are supported, each suchmodulation scheme typically uses symbols transmitting a different numberof bits per symbols (though some may happen to use the same number),Condition #D1-6 Condition #D1-7, and Condition #D1-8 may not besatisfied for some modulation schemes. In such a case, the followingconditions apply instead of Condition #D1-6 Condition #D1-7, andCondition #D1-8.

(Condition #D1-9)

The difference between K_(a) and K_(b) satisfies 0 or 1. That is,|K_(a)−K_(b)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (adenotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

(Condition #D1-10)

The difference between K_(a,1) and K_(b,1) satisfies 0 or 1. That is,|K_(a,1)−K_(b,1)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

(Condition #D1-11)

The difference between K_(a,2) and K_(b,2) satisfies 0 or 1. That is,|K_(a,2)−K_(b,2)| satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer thatsatisfies 0≤b≤N−1), a≠b)

As described above, bias among the phases being used to transmit thecoded blocks is removed by creating a relationship between the codedblock and the phase of multiplication. As such, data reception qualitymay be improved for the reception device.

As described above, N phase changing values (or phase changing sets) areneeded in order to perform a change of phase having a period (cycle) ofN with the scheme for the regular change of phase. As such, N phasechanging values (or phase changing sets) PHASE[0], PHASE[1], PHASE[2] .. . PHASE[N−2], and PHASE[N−1] are prepared. However, schemes exist forordering the phases in the stated order with respect to the frequencydomain. No limitation is intended in this regard. The N phase changingvalues (or phase changing sets) PHASE[0], PHASE[1], PHASE[2] . . .PHASE[N−2], and PHASE[N−1] may also change the phases of blocks in thetime domain or in the time-frequency domain to obtain a symbolarrangement. Although the above examples discuss a phase changing schemewith a period (cycle) of N, the same effects are obtainable using Nphase changing values (or phase changing sets) at random. That is, the Nphase changing values (or phase changing sets) need not always haveregular periodicity. As long as the above-described conditions aresatisfied, great quality data reception improvements are realizable forthe reception device.

Furthermore, given the existence of modes for spatial multiplexing MIMOschemes, MIMO schemes using a fixed precoding matrix, space-time blockcoding schemes, single-stream transmission, and schemes using a regularchange of phase, the transmission device (broadcaster, base station) mayselect any one of these transmission schemes.

As described in Non-Patent Literature 3, spatial multiplexing MIMOschemes involve transmitting signals s1 and s2, which are mapped using aselected modulation scheme, on each of two different antennas. MIMOschemes using a fixed precoding matrix involve performing precoding only(with no change in phase). Further, space-time block coding schemes aredescribed in Non-Patent Literature 9, 16, and 17. Single-streamtransmission schemes involve transmitting signal s1, mapped with aselected modulation scheme, from an antenna after performingpredetermined processing.

Schemes using multi-carrier transmission such as OFDM involve a firstcarrier group made up of a plurality of carriers and a second carriergroup made up of a plurality of carriers different from the firstcarrier group, and so on, such that multi-carrier transmission isrealized with a plurality of carrier groups. For each carrier group, anyof spatial multiplexing MIMO schemes, MIMO schemes using a fixedprecoding matrix, space-time block coding schemes, single-streamtransmission, and schemes using a regular change of phase may be used.In particular, schemes using a regular change of phase on a selected(sub-)carrier group are preferably used to realize the above.

Although the present description describes the present Embodiment as atransmission device applying precoding, baseband switching, and changein phase, all of these may be variously combined. In particular, thephase changer discussed for the present Embodiment may be freelycombined with the change in phase discussed in all other Embodiments.

Embodiment D2

The present Embodiment describes a phase change initialization schemefor the regular change of phase described throughout the presentdescription. This initialization scheme is applicable to thetransmission device from FIG. 4 when using a multi-carrier scheme suchas OFDM, and to the transmission devices of FIGS. 67 and 70 when using asingle encoder and distributor, similar to FIG. 4.

The following is also applicable to a scheme for regularly changing thephase when encoding is performed using block codes as described inNon-Patent Literature 12 through 15, such as QC LDPC Codes (not onlyQC-LDPC but also LDPC codes may be used), concatenated LDPC and BCHcodes, Turbo codes or Duo-Binary Turbo Codes using tail-biting, and soon.

The following example considers a case where two streams s1 and s2 aretransmitted. When encoding has been performed using block codes andcontrol information and the like is not necessary, the number of bitsmaking up each coded block matches the number of bits making up eachblock code (control information and so on described below may yet beincluded). When encoding has been performed using block codes or thelike and control information or the like (e.g., CRC transmissionparameters) is required, then the number of bits making up each codedblock is the sum of the number of bits making up the block codes and thenumber of bits making up the information.

FIG. 34 illustrates the varying numbers of symbols and slots needed ineach coded block when block codes are used. FIG. 34 illustrates thevarying numbers of symbols and slots needed in each coded block whenblock codes are used when, for example, two streams s1 and s2 aretransmitted as indicated by the above-described transmission device, andthe transmission device has only one encoder. (Here, the transmissionscheme may be any single-carrier scheme or multi-carrier scheme such asOFDM.)

As shown in FIG. 34, when block codes are used, there are 6000 bitsmaking up a single coded block. In order to transmit these 6000 bits,the number of required symbols depends on the modulation scheme, being3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM.

Then, given that the above-described transmission device transmits twostreams simultaneously, 1500 of the aforementioned 3000 symbols neededwhen the modulation scheme is QPSK are assigned to s1 and the other 1500symbols are assigned to s2. As such, 1500 slots for transmitting the1500 symbols (hereinafter, slots) are required for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 slotsare needed to transmit all of the bits making up each coded block, andwhen the modulation scheme is 64-QAM, 500 slots are needed to transmitall of the bits making up each coded block.

The following describes a transmission device transmitting modulatedsignals having a frame configuration illustrated by FIGS. 71A and 71B.FIG. 71A illustrates a frame configuration for modulated signal z1′ orz1 (transmitted by antenna 312A) in the time and frequency domains.Similarly, FIG. 71B illustrates a frame configuration for modulatedsignal z2 (transmitted by antenna 312B) in the time and frequencydomains. Here, the frequency (band) used by modulated signal z1′ or z1and the frequency (band) used for modulated signal z2 are identical,carrying modulated signals z1′ or z1 and z2 at the same time.

As shown in FIG. 71A, the transmission device transmits a preamble(control symbol) during interval A. The preamble is a symboltransmitting control information for another party. In particular, thispreamble includes information on the modulation scheme used to transmita first and a second coded block. The transmission device transmits thefirst coded block during interval B. The transmission device thentransmits the second coded block during interval C.

Further, the transmission device transmits a preamble (control symbol)during interval D. The preamble is a symbol transmitting controlinformation for another party. In particular, this preamble includesinformation on the modulation scheme used to transmit a third or fourthcoded block and so on. The transmission device transmits the third codedblock during interval E. The transmission device then transmits thefourth coded block during interval D.

Also, as shown in FIG. 71B, the transmission device transmits a preamble(control symbol) during interval A. The preamble is a symboltransmitting control information for another party. In particular, thispreamble includes information on the modulation scheme used to transmita first and a second coded block. The transmission device transmits thefirst coded block during interval B. The transmission device thentransmits the second coded block during interval C.

Further, the transmission device transmits a preamble (control symbol)during interval D. The preamble is a symbol transmitting controlinformation for another party. In particular, this preamble includesinformation on the modulation scheme used to transmit a third or fourthcoded block and so on. The transmission device transmits the third codedblock during interval E. The transmission device then transmits thefourth coded block during interval D.

FIG. 72 indicates the number of slots used when transmitting the codedblocks from FIG. 34, specifically using 16-QAM as the modulation schemefor the first coded block. Here, 750 slots are needed to transmit thefirst coded block. Similarly, FIG. 72 also indicates the number of slotsused to transmit the second coded block, using QPSK as the modulationscheme therefor. Here, 1500 slots are needed to transmit the secondcoded block.

FIG. 73 indicates the slots used when transmitting the coded blocks fromFIG. 34, specifically using QPSK as the modulation scheme for the thirdcoded block. Here, 1500 slots are needed to transmit the coded block.

As explained throughout this description, modulated signal z1, i.e., themodulated signal transmitted by antenna 312A, does not undergo a changein phase, while modulated signal z2, i.e., the modulated signaltransmitted by antenna 312B, does undergo a change in phase. Thefollowing phase changing scheme is used for FIGS. 72 and 73.

Before the change in phase can occur, seven different phase changingvalues is prepared. The seven phase changing values are labeled #0, #1,#2, #3, #4, #5, #6, and #7. The change in phase is regular and periodic.In other words, the phase changing values are applied regularly andperiodically, such that the order is #0, #1, #2, #3, #4, #5, #6, #0, #1,#2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6 and so on.

As shown in FIG. 72, given that 750 slots are needed for the first codedblock, phase changing value #0 is used initially, such that #0, #1, #2,#3, #4, #5, #6, #0, #1, #2 . . . #3, #4, #5, #6 are used in succession,with the 750th slot using #0 at the final position.

The change in phase is then applied to each slot for the second codedblock. The present description assumes multi-cast transmission andbroadcasting applications. As such, a receiving terminal may have noneed for the first coded block and extract only the second coded block.In such circumstances, given that the final slot used for the firstcoded block uses phase changing value #0, the initial phase changingvalue used for the second coded block is #1. As such, the followingschemes are conceivable:

(a): The aforementioned terminal monitors the transmission of the firstcoded block, i.e., monitors the pattern of the phase changing valuesthrough the final slot used to transmit the first coded block, and thenestimates the phase changing value used for the initial slot of thesecond coded block;

(b): (a) does not occur, and the transmission device transmitsinformation on the phase changing values in use at the initial slot ofthe second coded block. Scheme (a) leads to greater energy consumptionby the terminal due to the need to monitor the transmission of the firstcoded block. However, scheme (b) leads to reduced data transmissionefficiency.

Accordingly, there is a need to improve the phase changing valueallocation described above. Consider a scheme in which the phasechanging value used to transmit the initial slot of each coded block isfixed. Thus, as indicated in FIG. 72, the phase changing value used totransmit the initial slot of the second coded block and the phasechanging value used to transmit the initial slot of the first codedblock are identical, being #0.

Similarly, as indicated in FIG. 73, the phase changing value used totransmit the initial slot of the third coded block is not #3, but isinstead identical to the phase changing value used to transmit theinitial slot of the first and second coded blocks, being #0.

As such, the problems accompanying both schemes (a) and (b) describedabove can be constrained while retaining the effects thereof.

In the present Embodiment, the scheme used to initialize the phasechanging value for each coded block, i.e., the phase changing value usedfor the initial slot of each coded block, is fixed so as to be #0.However, other schemes may also be used for single-frame units. Forexample, the phase changing value used for the initial slot of a symboltransmitting information after the preamble or control symbol has beentransmitted may be fixed at #0.

Embodiment D3

The above-described Embodiments discuss a weighting unit using aprecoding matrix expressed in complex numbers for precoding. However,the precoding matrix may also be expressed in real numbers.

That is, suppose that two baseband signals s1(i) and s2(i) (where i istime or frequency) have been mapped (using a modulation scheme), andprecoded to obtained precoded baseband signals z1(i) and z2(i). As such,mapped baseband signal s1(i) has an in-phase component of I_(s1)(i) anda quadrature component of Q_(s1)(i), and mapped baseband signal s2(i)has an in-phase component of I_(s2)(i) and a quadrature component ofQ_(s2)(i), while precoded baseband signal z1(i) has an in-phasecomponent of I_(z1)(i) and a quadrature component of Q_(z1)(i), andprecoded baseband signal z2(i) has an in-phase component of I_(z2)(i)and a quadrature component of Q_(z2)(i), which gives the followingprecoding matrix H_(r) when all values are real numbers.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 76} \right\rbrack & \; \\{\begin{pmatrix}{I_{z\; 1}(i)} \\{Q_{z\; 1}(i)} \\{I_{z\; 2}(i)} \\{Q_{z\; 2}(i)}\end{pmatrix} = {H_{r}\begin{pmatrix}{I_{s\; 1}(i)} \\{Q_{s\; 1}(i)} \\{I_{s\; 2}(i)} \\{Q_{s2}(i)}\end{pmatrix}}} & \left( {{formula}\mspace{14mu} 76} \right)\end{matrix}$

Precoding matrix H_(r) may also be expressed as follows, where allvalues are real numbers.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 77} \right\rbrack & \; \\{H_{r} = \begin{pmatrix}a_{11} & a_{12} & a_{13} & a_{14} \\a_{21} & a_{22} & a_{23} & a_{24} \\a_{31} & a_{32} & a_{33} & a_{34} \\a_{41} & a_{42} & a_{43} & a_{44}\end{pmatrix}} & \left( {{formula}\mspace{14mu} 77} \right)\end{matrix}$

where a₁₁, a₁₂, a₁₃, a₁₄, a₂₁, a₂₂, a₂₃, a₂₄, a₃₁, a₃₂, a₃₃, a₃₄, a₄₁,a₄₂, a₄₃, and a₄₄ are real numbers. However, none of the following mayhold: {a₁₁=0, a₁₂=0, a₁₃=0, and a₁₄=0}, {a₂₁=0, a₂₂=0, a₂₃=0, anda₂₄=0}, {a₃₁=0, a₃₂=0, a₃₃=0, and a₃₄=0}, and {a₄₁=0, a₄₂=0, a₄₃=0, anda₄₄=0}. Also, none of the following may hold: {a₁₁=0, a₂₁=0, a₃₁=0, anda₄₁=0}, {a₁₂=0, a₂₂=0, a₃₂=0, and a₄₂=0}, {a₁₃=0, a₂₃=0, a₃₃=0, anda₄₃=0}, and {a₁₄=0, a₂₄=0, a₃₄=0, and a₄₄=0}.

Embodiment E1

The present embodiment describes a scheme of initializing phase changein a case where (i) the transmission device in FIG. 4 is used, (ii) thetransmission device in FIG. 4 is compatible with the multi-carrierscheme such as the OFDM scheme, and (iii) one encoder and a distributoris adopted in the transmission device in FIG. 67 and the transmissiondevice in FIG. 70 as shown in FIG. 4, when the phase change scheme forregularly performing phase change described in this description is used.

The following describes the scheme for regularly changing the phase whenusing a Quasi-Cyclic Low-Density Parity-Check (QC-LDPC) code (or an LDPCcode other than a QC-LDPC code), a concatenated code consisting of anLDPC code and a Bose-Chaudhuri-Hocquenghem (BCH) code, and a block codesuch as a turbo code or a duo-binary turbo code using tail-biting. Thesecodes are described in Non-Patent Literatures 12 through 15.

The following describes a case of transmitting two streams s1 and s2 asan example. Note that, when the control information and the like are notrequired to perform encoding using the block code, the number of bitsconstituting the coding (encoded) block is the same as the number ofbits constituting the block code (however, the control information andthe like described below may be included). When the control informationand the like (e.g. CRC (cyclic redundancy check), a transmissionparameter) are required to perform encoding using the block code, thenumber of bits constituting the coding (encoded) block can be a sum ofthe number of bits constituting the block code and the number of bits ofthe control information and the like.

FIG. 34 shows a change in the number of symbols and slots required forone coding (encoded) block when the block code is used. FIG. 34 shows achange in the number of symbols and slots required for one coding(encoded) block when the block code is used in a case where the twostreams s1 and s2 are transmitted and the transmission device has asingle encoder, as shown in the transmission device described above(note that, in this case, either the single carrier transmission or themulti-carrier transmission such as the OFDM may be used as atransmission system).

As shown in FIG. 34, let the number of bits constituting one coding(encoded) block in the block code be 6000 bits. In order to transmit the6000 bits, 3000 symbols, 1500 symbols and 1000 symbols are necessarywhen the modulation scheme is QPSK, 16QAM and 64QAM, respectively.

Since two streams are to be simultaneously transmitted in thetransmission device above, when the modulation scheme is QPSK, 1500symbols are allocated to s1 and remaining 1500 symbols are allocated tos2 out of the above-mentioned 3000 symbols. Therefore, 1500 slots(referred to as slots) are necessary to transmit 1500 symbols by s1 andtransmit 1500 symbols by s2.

Making the same considerations, 750 slots are necessary to transmit allthe bits constituting one coding (encoded) block when the modulationscheme is 16QAM, and 500 slots are necessary to transmit all the bitsconstituting one block when the modulation scheme is 64QAM.

Next, a case where the transmission device transmits modulated signalseach having a frame structure shown in FIGS. 71A and 71B is considered.FIG. 71A shows a frame structure in the time and frequency domain for amodulated signal z′1 or z1 (transmitted by the antenna 312A). FIG. 71Bshows a frame structure in the time and frequency domain for a modulatedsignal z2 (transmitted by the antenna 312B). In this case, the modulatedsignal z′1 or z1 and the modulated signal z2 are assumed to occupy thesame frequency (bandwidth), and the modulated signal z′1 or z1 and themodulated signal z2 are assumed to exist at the same time.

As shown in FIG. 71A, the transmission device transmits a preamble(control symbol) in an interval A. The preamble is a symbol fortransmitting control information to the communication partner and isassumed to include information on the modulation scheme for transmittingthe first coding (encoded) block and the second coding (encoded) block.The transmission device is to transmit the first coding (encoded) blockin an interval B. The transmission device is to transmit the secondcoding (encoded) block in an interval C.

The transmission device transmits the preamble (control symbol) in aninterval D. The preamble is a symbol for transmitting controlinformation to the communication partner and is assumed to includeinformation on the modulation scheme for transmitting the third coding(encoded) block, the fourth coding (encoded) block and so on. Thetransmission device is to transmit the third coding (encoded) block inan interval E. The transmission device is to transmit the fourth coding(encoded) block in an interval F.

As shown in FIG. 71B, the transmission device transmits a preamble(control symbol) in the interval A. The preamble is a symbol fortransmitting control information to the communication partner and isassumed to include information on the modulation scheme for transmittingthe first coding (encoded) block and the second coding (encoded) block.The transmission device is to transmit the first coding (encoded) blockin the interval B. The transmission device is to transmit the secondcoding (encoded) block in the interval C.

The transmission device transmits the preamble (control symbol) in theinterval D. The preamble is a symbol for transmitting controlinformation to the communication partner and is assumed to includeinformation on the modulation scheme for transmitting the third coding(encoded) block, the fourth coding (encoded) block and so on. Thetransmission device is to transmit the third coding (encoded) block inthe interval E. The transmission device is to transmit the fourth coding(encoded) block in the interval F.

FIG. 72 shows the number of slots used when the coding (encoded) blocksare transmitted as shown in FIG. 34, and, in particular, when 16QAM isused as the modulation scheme in the first coding (encoded) block. Inorder to transmit first coding (encoded) block, 750 slots are necessary.

Similarly, FIG. 100 shows the number of slots used when QPSK is used asthe modulation scheme in the second coding (encoded) block. In order totransmit second coding (encoded) block, 1500 slots are necessary.

FIG. 73 shows the number of slots used when the coding (encoded) blockis transmitted as shown in FIG. 34, and, in particular, when QPSK isused as the modulation scheme in the third coding (encoded) block. Inorder to transmit third coding (encoded) block, 1500 slots arenecessary.

As described in this description, a case where phase change is notperformed for the modulated signal z1, i.e. the modulated signaltransmitted by the antenna 312A, and is performed for the modulatedsignal z2, i.e. the modulated signal transmitted by the antenna 312B, isconsidered. In this case, FIGS. 72 and 73 show the scheme of performingphase change.

First, assume that seven different phase changing values are prepared toperform phase change, and are referred to as #0, #1, #2, #3, #4, #5 and#6. The phase changing values are to be regularly and cyclically used.That is to say, the phase changing values are to be regularly andcyclically changed in the order such as #0, #1, #2, #3, #4, #5, #6, #0,#1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, . . . .

First, as shown in FIG. 72, 750 slots exist in the first coding(encoded) block. Therefore, starting from #0, the phase changing valuesare arranged in the order #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, . . ., #4, #5, #6, #0, and end using #0 for the 750^(th) slot.

Next, the phase changing values are to be applied to each slot in thesecond coding (encoded) block. Since this description is on theassumption that the phase changing values are applied to the multicastcommunication and broadcast, one possibility is that a receptionterminal does not need the first coding (encoded) block and extractsonly the second coding (encoded) block. In such a case, even when phasechanging value #0 is used to transmit the last slot in the first coding(encoded) block, the phase changing value #1 is used first to transmitthe second coding (encoded) block. In this case, the following twoschemes are considered:

(a) The above-mentioned terminal monitors how the first coding (encoded)block is transmitted, i.e. the terminal monitors a pattern of the phasechanging value used to transmit the last slot in the first coding(encoded) block, and estimates the phase changing value to be used totransmit the first slot in the second coding (encoded) block; and

(b) The transmission device transmits information on the phase changingvalue used to transmit the first slot in the second coding (encoded)block without performing (a).

In the case of (a), since the terminal has to monitor transmission ofthe first coding (encoded) block, power consumption increases. In thecase of (b), transmission efficiency of data is reduced.

Therefore, there is room for improvement in allocation of precodingmatrices as described above. In order to address the above-mentionedproblems, a scheme of fixing the phase changing value used to transmitthe first slot in each coding (encoded) block is proposed. Therefore, asshown in FIG. 72, the phase changing value used to transmit the firstslot in the second coding (encoded) block is set to #0 as with the phasechanging value used to transmit the first slot in the first coding(encoded) block.

Similarly, as shown in FIG. 73, the phase changing value used totransmit the first slot in the third coding (encoded) block is set notto #3 but to #0 as with the phase changing value used to transmit thefirst slot in the first coding (encoded) block and in the second coding(encoded) block.

With the above-mentioned scheme, an effect of suppressing the problemsoccurring in (a) and (b) is obtained.

Note that, in the present embodiment, the scheme of initializing thephase changing values in each coding (encoded) block, i.e. the scheme inwhich the phase changing value used to transmit the first slot in eachcoding (encoded) block is fixed to #0, is described. As a differentscheme, however, the phase changing values may be initialized in unitsof frames. For example, in the symbol for transmitting the preamble andinformation after transmission of the control symbol, the phase changingvalue used in the first slot may be fixed to #0.

For example, in FIG. 71, a frame is interpreted as starting from thepreamble, the first coding (encoded) block in the first frame is firstcoding (encoded) block, and the first coding (encoded) block in thesecond frame is the third coding (encoded) block. This exemplifies acase where “the phase changing value used in the first slot may be fixed(to #0) in units of frames” as described above using FIGS. 72 and 73.

The following describes a case where the above-mentioned scheme isapplied to a broadcasting system that uses the DVB-T2 standard. First,the frame structure for a broadcast system according to the DVB-T2standard is described.

FIG. 74 is an overview of the frame structure of a signal a signaltransmitted by a broadcast station according to the DVB-T2 standard.According to the DVB-T2 standard, an OFDM scheme is employed. Thus,frames are structured in the time and frequency domains. FIG. 74 showsthe frame structure in the time and frequency domains. The frame iscomposed of P1 Signalling data (7401), L1 Pre-Signalling data (7402), L1Post-Signalling data (7403), Common PLP (7404), and PLPs #1 to #N(7405_1 to 7405_N) (PLP: Physical Layer Pipe). (Here, L1 Pre-Signallingdata (7402) and L1 Post-Signalling data (7403) are referred to as P2symbols.) As above, the frame composed of P1 Signalling data (7401), L1Pre-Signalling data (7402), L1 Post-Signalling data (7403), Common PLP(7404), and PLPs #1 to #N (7405_1 to 7405_N) is referred to as a T2frame, which is a unit of frame structure.

The P1 Signalling data (7401) is a symbol for use by a reception devicefor signal detection and frequency synchronization (including frequencyoffset estimation). Also, the P1 Signalling data (7401) transmitsinformation including information indicating the FFT (Fast FourierTransform) size, and information indicating which of SISO (Single-InputSingle-Output) and MISO (Multiple-Input Single-Output) is employed totransmit a modulated signal. (The SISO scheme is for transmitting onemodulated signal, whereas the MISO scheme is for transmitting aplurality of modulated signals using space-time block codes shown inNon-Patent Literatures 9, 16 and 17.)

The L1 Pre-Signalling data (7402) transmits information including:information about the guard interval used in transmitted frames;information about the signal processing method for reducing PAPR (Peakto Average Power Ratio); information about the modulation scheme, errorcorrection scheme (FEC: Forward Error Correction), and coding rate ofthe error correction scheme all used in transmitting L1 Post-Signallingdata; information about the size of L1 Post-Signalling data and theinformation size; information about the pilot pattern; information aboutthe cell (frequency region) unique number; and information indicatingwhich of the normal mode and extended mode (the respective modes differsin the number of subcarriers used in data transmission) is used.

The L1 Post-Signalling data (7403) transmits information including:information about the number of PLPs; information about the frequencyregion used; information about the unique number of each PLP;information about the modulation scheme, error correction scheme, codingrate of the error correction scheme all used in transmitting the PLPs;and information about the number of blocks transmitted in each PLP.

The Common PLP (7404) and PLPs #1 to #N (7405_1 to 7405_N) are fieldsused for transmitting data.

In the frame structure shown in FIG. 74, the P1 Signalling data (7401),L1 Pre-Signalling data (7402), L1 Post-Signalling data (7403), CommonPLP (7404), and PLPs #1 to #N (7405_1 to 7405_N) are illustrated asbeing transmitted by time-sharing. In practice, however, two or more ofthe signals are concurrently present. FIG. 75 shows such an example. Asshown in FIG. 75, L1 Pre-Signalling data, L1 Post-Signalling data, andCommon PLP may be present at the same time, and PLP #1 and PLP #2 may bepresent at the same time. That is, the signals constitute a frame usingboth time-sharing and frequency-sharing.

FIG. 76 shows an example of the structure of a transmission deviceobtained by applying the phase change schemes of performing phase changeon the signal after performing precoding (or after performing precoding,and switching the baseband signals) to a transmission device compliantwith the DVB-T2 standard (i.e., to a transmission device of a broadcaststation). A PLP signal generator 7602 receives PLP transmission data(transmission data for a plurality of PLPs) 7601 and a control signal7609 as input, performs mapping of each PLP according to the errorcorrection scheme and modulation scheme indicated for the PLP by theinformation included in the control signal 7609, and outputs a(quadrature) baseband signal 7603 carrying a plurality of PLPs.

A P2 symbol signal generator 7605 receives P2 symbol transmission data7604 and the control signal 7609 as input, performs mapping according tothe error correction scheme and modulation scheme indicated for each P2symbol by the information included in the control signal 7609, andoutputs a (quadrature) baseband signal 7606 carrying the P2 symbols.

A control signal generator 7608 receives P1 symbol transmission data7607 and P2 symbol transmission data 7604 as input, and then outputs, asthe control signal 7609, information about the transmission scheme (theerror correction scheme, coding rate of the error correction, modulationscheme, block length, frame structure, selected transmission schemesincluding a transmission scheme that regularly hops between precodingmatrices, pilot symbol insertion scheme, IFFT (Inverse Fast FourierTransform)/FFT, method of reducing PAPR, and guard interval insertionscheme) of each symbol group shown in FIG. 74 (P1 Signalling data(7401), L1 Pre-Signalling data (7402), L1 Post-Signalling data (7403),Common PLP (7404), PLPs #1 to #N (7405_1 to 7405_N)).

A frame configurator 7610 receives, as input, the baseband signal 7603carrying PLPs, the baseband signal 7606 carrying P2 symbols, and thecontrol signal 7609. On receipt of the input, the frame configurator7610 changes the order of input data in frequency domain and time domainbased on the information about frame structure included in the controlsignal, and outputs a (quadrature) baseband signal 7611_1 correspondingto stream 1 (a signal after the mapping, that is, a baseband signalbased on a modulation scheme to be used) and a (quadrature) basebandsignal 7611_2 corresponding to stream 2 (a signal after the mapping,that is, a baseband signal based on a modulation scheme to be used) bothin accordance with the frame structure.

A signal processor 7612 receives, as input, the baseband signal 7611_1corresponding to stream 1, the baseband signal 7611_2 corresponding tostream 2, and the control signal 7609 and outputs a modulated signal 1(7613_1) and a modulated signal 2 (7613_2) each obtained as a result ofsignal processing based on the transmission scheme indicated byinformation included in the control signal 7609.

The characteristic feature noted here lies in the following. That is,when a transmission scheme that performs phase change on the signalafter performing precoding (or after performing precoding, and switchingthe baseband signals) is selected, the signal processor performs phasechange on signals after performing precoding (or after performingprecoding, and switching the baseband signals) in a manner similar toFIGS. 6, 25, 26, 27, 28, 29 and 69. Thus, processed signals so obtainedare the modulated signal 1 (7613_1) and modulated signal 2 (7613_2)obtained as a result of the signal processing.

A pilot inserter 7614_1 receives, as input, the modulated signal 1(7613_1) obtained as a result of the signal processing and the controlsignal 7609, inserts pilot symbols into the received modulated signal 1(7613_1), and outputs a modulated signal 7615_1 obtained as a result ofthe pilot signal insertion. Note that the pilot symbol insertion iscarried out based on information indicating the pilot symbol insertionscheme included the control signal 7609.

A pilot inserter 7614_2 receives, as input, the modulated signal 2(76132) obtained as a result of the signal processing and the controlsignal 7609, inserts pilot symbols into the received modulated signal 2(76132), and outputs a modulated signal 7615_2 obtained as a result ofthe pilot symbol insertion. Note that the pilot symbol insertion iscarried out based on information indicating the pilot symbol insertionscheme included the control signal 7609.

An IFFT (Inverse Fast Fourier Transform) unit 7616_1 receives, as input,the modulated signal 7615_1 obtained as a result of the pilot symbolinsertion and the control signal 7609, and applies IFFT based on theinformation about the IFFT method included in the control signal 7609,and outputs a signal 7617_1 obtained as a result of the IFFT.

An IFFT unit 7616_2 receives, as input, the modulated signal 7615_2obtained as a result of the pilot symbol insertion and the controlsignal 7609, and applies IFFT based on the information about the IFFTmethod included in the control signal 7609, and outputs a signal 7617_2obtained as a result of the IFFT.

A PAPR reducer 7618_1 receives, as input, the signal 7617_1 obtained asa result of the IFFT and the control signal 7609, performs processing toreduce PAPR on the received signal 7617_1, and outputs a signal 7619_1obtained as a result of the PAPR reduction processing. Note that thePAPR reduction processing is performed based on the information aboutthe PAPR reduction included in the control signal 7609.

A PAPR reducer 7618_2 receives, as input, the signal 76172 obtained as aresult of the IFFT and the control signal 7609, performs processing toreduce PAPR on the received signal 76172, and outputs a signal 7619_2obtained as a result of the PAPR reduction processing. Note that thePAPR reduction processing is carried out based on the information aboutthe PAPR reduction included in the control signal 7609.

A guard interval inserter 7620_1 receives, as input, the signal 7619_1obtained as a result of the PAPR reduction processing and the controlsignal 7609, inserts guard intervals into the received signal 7619_1,and outputs a signal 7621_1 obtained as a result of the guard intervalinsertion. Note that the guard interval insertion is carried out basedon the information about the guard interval insertion scheme included inthe control signal 7609.

A guard interval inserter 7620_2 receives, as input, the signal 7619_2obtained as a result of the PAPR reduction processing and the controlsignal 7609, inserts guard intervals into the received signal 76192, andoutputs a signal 7621_2 obtained as a result of the guard intervalinsertion. Note that the guard interval insertion is carried out basedon the information about the guard interval insertion scheme included inthe control signal 7609.

A P1 symbol inserter 7622 receives, as input, the signal 7621_1 obtainedas a result of the guard interval insertion, the signal 7621_2 obtainedas a result of the guard interval insertion, and the P1 symboltransmission data 7607, generates a P1 symbol signal from the P1 symboltransmission data 7607, adds the P1 symbol to the signal 7621_1 obtainedas a result of the guard interval insertion, and adds the P1 symbol tothe signal 7621_2 obtained as a result of the guard interval insertion.Then, the P1 symbol inserter 7622 outputs a signal 7623_1 as a result ofthe addition of the P1 symbol and a signal 7623_2 as a result of theaddition of the P1 symbol. Note that a P1 symbol signal may be added toboth the signals 7623_1 and 7623_2 or to one of the signals 7623_1 and7623_2. In the case where the P1 symbol signal is added to one of thesignals 7623_1 and 76232, the following is noted. For purposes ofdescription, an interval of the signal to which a P1 symbol is added isreferred to as a P1 symbol interval. Then, the signal to which a P1signal is not added includes, as a baseband signal, a zero signal in aninterval corresponding to the P1 symbol interval of the other signal.

A wireless processor 7624_1 receives the signal 7623_1 obtained as aresult of the processing related to P1 symbol and the control signal7609, performs processing such as frequency conversion, amplification,and the like, and outputs a transmission signal 7625_1. The transmissionsignal 7625_1 is then output as a radio wave from an antenna 7626_1.

A wireless processor 7624_2 receives the signal 7623_2 obtained as aresult of the processing related to P1 symbol and the control signal7609, performs processing such as frequency conversion, amplification,and the like, and outputs a transmission signal 7625_2. The transmissionsignal 7625_2 is then output as a radio wave from an antenna 7626_2.

As described above, by the P1 symbol, P2 symbol and control symbolgroup, information on transmission scheme of each PLP (for example, atransmission scheme of transmitting a single modulated signal, atransmission scheme of performing phase change on the signal afterperforming precoding (or after performing precoding, and switching thebaseband signals)) and a modulation scheme being used is transmitted toa terminal. In this case, if the terminal extracts only PLP that isnecessary as information to perform demodulation (including separationof signals and signal detection) and error correction decoding, powerconsumption of the terminal is reduced. Therefore, as described usingFIGS. 71 through 73, the scheme in which the phase changing value usedin the first slot in the PLP transmitted using, as the transmissionscheme, the transmission scheme for regularly performing phase change onthe signal after performing precoding (or after performing precoding,and switching the baseband signals) is fixed (to #0) is proposed. Notethat the PLP transmission scheme is not limited to those describedabove. For example, a transmission scheme using space-time block codesdisclosed in Non-Patent Literatures 9, 16 and 17 or another transmissionscheme may be adopted.

For example, assume that the broadcast station transmits each symbolhaving the frame structure as shown in FIG. 74. In this case, as anexample, FIG. 77 shows a frame structure in frequency-time domain whenthe broadcast station transmits PLP $1 (to avoid confusion, #1 isreplaced by $1) and PLP $K using the transmission scheme of performingphase change on the signal after performing precoding (or afterperforming precoding, and switching the baseband signals).

Note that, in the following description, as an example, assume thatseven phase changing values are prepared in the transmission scheme ofperforming phase change on the signal after performing precoding (orafter performing precoding, and switching the baseband signals), and arereferred to as #0, #1, #2, #3, #4, #5 and #6. The phase changing valuesare to be regularly and cyclically used. That is to say, the phasechanging values are to be regularly and cyclically changed in the ordersuch as #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, #0, #1,#2, #3, #4, #5, #6, . . . .

As shown in FIG. 77, the slot (symbol) in PLP $1 starts with a time Tand a carrier 3 (7701 in FIG. 77) and ends with a time T+4 and a carrier4 (7702 in FIG. 77) (see FIG. 77).

This is to say, in PLP $1, the first slot is the time T and the carrier3, the second slot is the time T and the carrier 4, the third slot isthe time T and a carrier 5, . . . , the seventh slot is a time T+1 and acarrier 1, the eighth slot is the time T+1 and a carrier 2, the ninthslot is the time T+1 and the carrier 3, . . . , the fourteenth slot isthe time T+1 and a carrier 8, the fifteenth slot is a time T+2 and acarrier 0, . . . .

The slot (symbol) in PLP $K starts with a time S and a carrier 4 (7703in FIG. 77) and ends with a time S+8 and the carrier 4 (7704 in FIG. 77)(see FIG. 77). This is to say, in PLP $K, the first slot is the time Sand the carrier 4, the second slot is the time S and a carrier 5, thethird slot is the time S and a carrier 6, . . . , the fifth slot is thetime S and a carrier 8, the ninth slot is a time S+1 and a carrier 1,the tenth slot is the time S+1 and a carrier 2 . . . , the sixteenthslot is the time S+1 and the carrier 8, the seventeenth slot is a timeS+2 and a carrier 0, . . . .

Note that information on slot that includes information on the firstslot (symbol) and the last slot (symbol) in each PLP and is used by eachPLP is transmitted by the control symbol including the P1 symbol, the P2symbol and the control symbol group.

In this case, as described using FIGS. 71 through 73, the first slot inPLP $1, which is the time T and the carrier 3 (7701 in FIG. 77), issubject to phase change using the phase changing value #0. Similarly,the first slot in PLP $K, which is the time S and the carrier 4 (7703 inFIG. 77), is subject tophase change using the phase changing value #0regardless of the number of the phase changing values used in the lastslot in PLP $K−1, which is the time S and the carrier 3 (7705 in FIG.77). (However, as described above, it is assumed that precoding (orswitching the precoding matrices and baseband signals) has beenperformed before the phase change is performed).

Also, the first slot in another PLP transmitted using a transmissionscheme that performs phase change on the signal after performingprecoding (or after performing precoding, and switching the basebandsignals) is precoded using the precoding matrix #0.

With the above-mentioned scheme, an effect of suppressing the problemsdescribed in Embodiment D2 above, occurring in (a) and (b) is obtained.

Naturally, the reception device extracts necessary PLP from theinformation on slot that is included in the control symbol including theP1 symbol, the P2 symbol and the control symbol group and is used byeach PLP to perform demodulation (including separation of signals andsignal detection) and error correction decoding. The reception devicelearns a phase change rule of regularly performing phase change on thesignal after performing precoding (or after performing precoding, andswitching the baseband signals) in advance (when there are a pluralityof rules, the transmission device transmits information on the rule tobe used, and the reception device learns the rule being used byobtaining the transmitted information). By synchronizing a timing ofrules of switching the phase changing values based on the number of thefirst slot in each PLP, the reception device can perform demodulation ofinformation symbols (including separation of signals and signaldetection).

Next, a case where the broadcast station (base station) transmits amodulated signal having a frame structure shown in FIG. 78 is considered(the frame composed of symbol groups shown in FIG. 78 is referred to asa main frame). In FIG. 78, elements that operate in a similar way toFIG. 74 bear the same reference signs. The characteristic feature isthat the main frame is separated into a subframe for transmitting asingle modulated signal and a subframe for transmitting a plurality ofmodulated signals so that gain control of received signals can easily beperformed. Note that the expression “transmitting a single modulatedsignal” also indicates that a plurality of modulated signals that arethe same as the single modulated signal transmitted from a singleantenna are generated, and the generated signals are transmitted fromrespective antennas.

In FIG. 78, PLP #1 (7405_1) through PLP #N (7405_N) constitute asubframe 7800 for transmitting a single modulated signal. The subframe7800 is composed only of PLPs, and does not include PLP for transmittinga plurality of modulated signals. Also, PLP $1 (7802_1) through PLP $M(7802_M) constitute a subframe 7801 for transmitting a plurality ofmodulated signals. The subframe 7801 is composed only of PLPs, and doesnot include PLP for transmitting a single modulated signal.

In this case, as described above, when the above-mentioned transmissionscheme for regularly performing phase change on the signal afterperforming precoding (or after performing precoding, and switching thebaseband signals) is used in the subframe 7801, the first slot in PLP(PLP $1 (7802_1) through PLP $M (7802_M)) is assumed to be precodedusing the precoding matrix #0 (referred to as initialization of theprecoding matrices). The above-mentioned initialization of precodingmatrices, however, is irrelevant to a PLP in which another transmissionscheme, for example, one of the transmission scheme not performing phasechange, the transmission scheme using the space-time block codes and thetransmission scheme using a spatial multiplexing MIMO system (see FIG.23) is used in PLP $1 (7802_1) through PLP $M (7802_M).

As shown in FIG. 79, PLP $1 is assumed to be the first PLP in thesubframe for transmitting a plurality of modulated signals in the Xthmain frame. Also, PLP $1′ is assumed to be the first PLP in the subframefor transmitting a plurality of modulated signals in the Yth main frame(Y is not X). Both PLP $1 and PLP $1′ are assumed to use thetransmission scheme for regularly performing phase change on the signalafter performing precoding (or after performing precoding, and switchingthe baseband signals). In FIG. 79, elements that operate in a similarway to FIG. 77 bear the same reference signs.

In this case, the first slot (7701 in FIG. 79 (time T and carrier 3)) inPLP $1, which is the first PLP in the subframe for transmitting aplurality of modulated signals in the Xth main frame, is assumed to besubject to phase change using the phase changing value #0.

Similarly, the first slot (7901 in FIG. 79 (time T′ and carrier 7)) inPLP $1′, which is the first PLP in the subframe for transmitting aplurality of modulated signals in the Yth main frame, is assumed to besubject to phase change using the phase changing value #0.

As described above, in each main frame, the first slot in the first PLPin the subframe for transmitting a plurality of modulated signals ischaracterized by being subject to phase change using the phase changingvalue #0.

This is also important to suppress the problems described in EmbodimentD2 occurring in (a) and (b).

Note that since the the first slot (7701 in FIG. 79 (time T and carrier3)) in PLP $1 is assumed to be subject to phase change using the phasechanging value #0, when the phase changing value is updated in thetime-frequency domain, the slot at time T, carrier 4 is subject to phasechange using the phase changing value #1, the slot at time T, carrier 5is subject to phase change using the phase changing value #2, the slotat time T, carrier 6 is subject to phase change using the phase changingvalue #3, and so on.

Similarly, note that since the first slot (7901 in FIG. 79 (time T′ andcarrier 7)) in PLP $1 is assumed to be subject to phase change using thephase changing value #0, when the phase changing value is updated in thetime-frequency domain, the slot at time T′, carrier 8 is subject tophase change using the phase changing value #1, the slot at time T′+1,carrier 1 is subject to phase change using the phase changing value #2,the slot at time T′+2, carrier 1 is subject to phase change using thephase changing value #3, the slot at time T′+3, carrier 1 is subject tophase change using the phase changing value #4, and so on.

Note that, in the present embodiment, cases where (i) the transmissiondevice in FIG. 4 is used, (ii) the transmission device in FIG. 4 iscompatible with the multi-carrier scheme such as the OFDM scheme, and(iii) one encoder and a distributor is adopted in the transmissiondevice in FIG. 67 and the transmission device in FIG. 70 as shown inFIG. 4 are taken as examples. The initialization of phase changingvalues described in the present embodiment, however, is also applicableto a case where the two streams s1 and s2 are transmitted and thetransmission device has two single encoders as shown in the transmissiondevice in FIG. 3, the transmission device in FIG. 12, the transmissiondevice in FIG. 67 and the transmission device in FIG. 70.

The transmission devices pertaining to the present invention, asillustrated by FIGS. 3, 4, 12, 13, 51, 52, 67, 70, 76, and so ontransmit two modulated signals, namely modulated signal #1 and modulatedsignal #2, on two different transmit antennas. The average transmissionpower of the modulated signals #1 and #2 may be set freely. For example,when the two modulated signals each have a different averagetransmission power, conventional transmission power control technologyused in wireless transmission systems may be applied thereto. Therefore,the average transmission power of modulated signals #1 and #2 maydiffer. In such circumstances, transmission power control may be appliedto the baseband signals (e.g., when mapping is performed using themodulation scheme), or may be performed by a power amplifier immediatelybefore the antenna.

Embodiment F1

The schemes for regularly performing phase change on the modulatedsignal after precoding described in Embodiments 1 through 4, EmbodimentA1, Embodiments C1 through C7, Embodiments D1 through D3 and EmbodimentE1 are applicable to any baseband signals s1 and s2 mapped in the IQplane. Therefore, in Embodiments 1 through 4, Embodiment A1, EmbodimentsC1 through C7, Embodiments D1 through D3 and Embodiment E1, the basebandsignals s1 and s2 have not been described in detail. On the other hand,when the scheme for regularly performing phase change on the modulatedsignal after precoding is applied to the baseband signals s1 and s2generated from the error correction coded data, excellent receptionquality can be achieved by controlling average power (average value) ofthe baseband signals s1 and s2. In the present embodiment, the followingdescribes a scheme of setting the average power of s1 and s2 when thescheme for regularly performing phase change on the modulated signalafter precoding is applied to the baseband signals s1 and s2 generatedfrom the error correction coded data.

As an example, the modulation schemes for the baseband signal s1 and thebaseband signal s2 are described as QPSK and 16QAM, respectively.

Since the modulation scheme for s1 is QPSK, s1 transmits two bits persymbol. Let the two bits to be transmitted be referred to as b0 and b1.On the other hand, since the modulation scheme for s2 is 16QAM, s2transmits four bits per symbol. Let the four bits to be transmitted bereferred to as b2, b3, b4 and and b5. The transmission device transmitsone slot composed of one symbol for s1 and one symbol for s2, i.e. sixbits b0, b1, b2, b3, b4 and b5 per slot.

For example, in FIG. 80 as an example of signal point layout in the IQplane for 16QAM, (b2, b3, b4, b5)=(0, 0, 0, 0) is mapped onto (I,Q)=(3×g, 3×g), (b2, b3, b4, b5)=(0, 0, 0, 1) is mapped onto (I, Q)=(3×g,1×g), (b2, b3, b4, b5)=(0, 0, 1, 0) is mapped onto (I, Q)=(1×g, 3×g),(b2, b3, b4, b5)=(0, 0, 1, 1) is mapped onto (I, Q)=(1×g, 1×g), (b2, b3,b4, b5)=(0, 1, 0, 0) is mapped onto (I, Q)=(3×g, −3×g), . . . , (b2, b3,b4, b5)=(1, 1, 1, 0) is mapped onto (I, Q)=(−1×g, −3×g), and (b2, b3,b4, b5)=(1, 1, 1, 1) is mapped onto (I, Q)=(−1×g, −1×g). Note that b2through b5 shown on the top right of FIG. 80 shows the bits and thearrangement of the numbers shown on the IQ plane.

Also, in FIG. 81 as an example of signal point layout in the IQ planefor QPSK, (b0, b1)=(0, 0) is mapped onto (I, Q)=(1×h, 1×h), (b0,b1)=(0, 1) is mapped onto (I, Q)=(1×h, −1×h), (b0, b1)=(1, 0) is mappedonto (I, Q)=(−1× h, 1×h), and (b0, b1)=(1, 1) is mapped onto (I,Q)=(−1×h, −1×h). Note that b0 and b1 shown on the top right of FIG. 81shows the bits and the arrangement of the numbers shown on the IQ plane.

Here, assume that the average power of s1 is equal to the average powerof s2, i.e. h shown in FIG. 81 is represented by formula 78 and g shownin FIG. 80 is represented by formula 79.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 78} \right\rbrack & \; \\{h = \frac{z}{\sqrt{2}}} & \left( {{formula}\mspace{14mu} 78} \right) \\\left\lbrack {{Math}.\mspace{14mu} 79} \right\rbrack & \; \\{g = \frac{z}{\sqrt{10}}} & \left( {{Formula}\mspace{14mu} 79} \right)\end{matrix}$

FIG. 82 shows the log-likelihood ratio obtained by the reception devicein this case. FIG. 82 schematically shows absolute values of thelog-likelihood ratio for b0 through b5 described above when thereception device obtains the log-likelihood ratio. In FIG. 82, 8200 isthe absolute value of the log-likelihood ratio for b0, 8201 is theabsolute value of the log-likelihood ratio for b1, 8202 is the absolutevalue of the log-likelihood ratio for b2, 8203 is the absolute value ofthe log-likelihood ratio for b3, 8204 is the absolute value of thelog-likelihood ratio for b4, and 8205 is the absolute value of thelog-likelihood ratio for b5. In this case, as shown in FIG. 82, when theabsolute values of the log-likelihood ratio for b0 and b1 transmitted inQPSK are compared with the absolute values of the log-likelihood ratiofor b2 through b5 transmitted in 16QAM, the absolute values of thelog-likelihood ratio for b0 and b1 are higher than the absolute valuesof the log-likelihood ratio for b2 through b5. That is, reliability ofb0 and b1 in the reception device is higher than the reliability of b2through b5 in the reception device. This is because of the followingreason. When h is represented by formula 79 in FIG. 80, a minimumEuclidian distance between signal points in the IQ plane for QPSK is asfollows.[Math. 80]√{square root over (2)}z  (formula 80)

On the other hand, when h is represented by formula 78 in FIG. 78, aminimum Euclidian distance between signal points in the IQ plane for16QAM is as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 81} \right\rbrack & \; \\{\frac{2}{\sqrt{10}}z} & \left( {{Formula}\mspace{14mu} 81} \right)\end{matrix}$

If the reception device performs error correction decoding (e.g. beliefpropagation decoding such as a sum-product decoding in a case where thecommunication system uses LDPC codes) under this situation, due to adifference in reliability that “the absolute values of thelog-likelihood ratio for b0 and b1 are higher than the absolute valuesof the log-likelihood ratio for b2 through b5”, a problem that the datareception quality degrades in the reception device by being affected bythe absolute values of the log-likelihood ratio for b2 through b5arises.

In order to overcome the problem, the difference between the absolutevalues of the log-likelihood ratio for b0 and b1 and the absolute valuesof the log-likelihood ratio for b2 through b5 should be reduced comparedwith FIG. 82, as shown in FIG. 83.

Therefore, it is considered that the average power (average value) of s1is made to be different from the average power (average value) of s2.FIGS. 84 and 85 each show an example of the structure of the signalprocessor relating to a power changer (although being referred to as thepower changer here, the power changer may be referred to as an amplitudechanger or a weight unit) and the weighting (precoding) unit. In FIG.84, elements that operate in a similar way to FIG. 3 and FIG. 6 bear thesame reference signs. Also, in FIG. 85, elements that operate in asimilar way to FIG. 3, FIG. 6 and FIG. 84 bear the same reference signs.

The following explains some examples of operations of the power changer.

Example 1

First, an example of the operation is described using FIG. 84. Let s1(t)be the (mapped) baseband signal for the modulation scheme QPSK. Themapping scheme for s1(t) is as shown in FIG. 81, and h is as representedby formula 78. Also, let s2(t) be the (mapped) baseband signal for themodulation scheme 16QAM. The mapping scheme for s2(t) is as shown inFIG. 80, and g is as represented by formula 79. Note that t is time. Inthe present embodiment, description is made taking the time domain as anexample.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 16QAM by u.Let u be a real number, and u>1.0. Letting the precoding matrix used inthe scheme for regularly performing phase change on the modulated signalafter precoding be F and the phase changing value used for regularlyperforming phase change be y(t) (y(t) may be imaginary number having theabsolute value of 1, i.e. ej^(θ(t)), the following formula is satisfied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 82} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}e^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}1 & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu} 82} \right)\end{matrix}$

Therefore, a ratio of the average power for QPSK to the average powerfor 16QAM is set to 1:u². With this structure, the reception device isin a reception condition in which the absolute value of thelog-likelihood ratio shown in FIG. 83 is obtained. Therefore, datareception quality is improved in the reception device.

The following describes a case where u in the ratio of the average powerfor QPSK to the average power for 16QAM 1:u² is set as shown in thefollowing formula.[Math. 83]u=√{square root over (5)}  (formula 83)

In this case, the minimum Euclidian distance between signal points inthe IQ plane for QPSK and the minimum Euclidian distance between signalpoints in the IQ plane for 16QAM can be the same. Therefore, excellentreception quality can be achieved.

The condition that the minimum Euclidian distances between signal pointsin the IQ plane for two different modulation schemes are equalized,however, is a mere example of the scheme of setting the ratio of theaverage power for QPSK to the average power for 16QAM. For example,according to other conditions such as a code length and a coding rate ofan error correction code used for error correction codes, excellentreception quality may be achieved when the value u for power change isset to a value (higher value or lower value) different from the value atwhich the minimum Euclidian distances between signal points in the IQplane for two different modulation schemes are equalized. In order toincrease the minimum distance between candidate signal points obtainedat the time of reception, a scheme of setting the value u as shown inthe following formula is considered, for example.[Math. 84]u=√{square root over (2)}  (formula 84)

The value, however, is set appropriately according to conditionsrequired as a system. This will be described later in detail.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the value u for power change is set based onthe control signal (8400). The following describes setting of the valueu for power change based on the control signal (8400) in order toimprove data reception quality in the reception device in detail.

Example 1-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction coding used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-1 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected block lengthindicated by the control signal (8400). Here, a value for power changeset according to a block length X is referred to as u_(LX)

For example, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000). Inthis case, for example, by setting u_(L1000), u_(L1500) and u_(L3000) soas to be different from one another, a high error correction capabilitycan be achieved for each code length. Depending on the set code length,however, the effect might not be obtained even if the value for powerchange is changed. In such a case, even when the code length is changed,it is unnecessary to change the value for power change (for example,u_(L1000)=u_(L1500) may be satisfied. What is important is that two ormore values exist in u_(L1000), u_(L1500) and u_(L3000)).

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more code lengths that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the code length is set, and performspower change.

Example 1-2

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a coding rate for the error correctioncodes used to generate s1 and s2 when the transmission device supports aplurality of coding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-2 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected coding rateindicated by the control signal (8400). Here, a value for power changeset according to a coding rate rx is referred to as u_(rX).

For example, when r1 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r1). When r2 is selected asthe coding rate, the power changer (8401B) sets a value for power changeto u_(r2). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3). In this case, forexample, by setting u_(r1), u_(r2) and u_(r3) so as to be different fromone another, a high error correction capability can be achieved for eachcoding rate. Depending on the set coding rate, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the coding rate is changed, it is unnecessary to changethe value for power change (for example, u_(r1)=u_(r2) may be satisfied.What is important is that two or more values exist in u_(r1), u_(r2) andu_(r3)).

Note that, as examples of r1, r2 and r3 described above, coding rates1/2, 2/3 and 3/4 are considered when the error correction code is theLDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more coding rates that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the coding rate is set, and performspower change.

Example 1-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto QPSK and the modulation scheme for s2 is changed from 16QAM to 64QAMby the control signal (or can be set to either 16QAM or 64QAM) isconsidered. Note that, in a case where the modulation scheme for s2(t)is 64QAM, the mapping scheme for s2(t) is as shown in FIG. 86. In FIG.86, k is represented by the following formula.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 85} \right\rbrack & \; \\{k = \frac{z}{\sqrt{42}}} & \left( {{formula}\mspace{14mu} 85} \right)\end{matrix}$

By performing mapping in this way, the average power obtained when h inFIG. 81 for QPSK is represented by formula 78 becomes equal to theaverage power obtained when g in FIG. 80 for 16QAM is represented byformula 79. In the mapping in 64QAM, the values I and Q are determinedfrom an input of six bits. In this regard, the mapping 64QAM may beperformed similarly to the mapping in QPSK and 16QAM.

That is to say, in FIG. 86 as an example of signal point layout in theIQ plane for 64QAM, (b0, b1, b2, b3, b4, b5)=(0, 0, 0, 0, 0, 0) ismapped onto (I, Q)=(7×k, 7×k), (b0, b1, b2, b3, b4, b5)=(0, 0, 0, 0,0, 1) is mapped onto (I, Q)=(7×k, 5×k), (b0, b1, b2, b3, b4, b5)=(0, 0,0, 0, 1, 0) is mapped onto (I, Q)=(5×k, 7×k), (b0, b1, b2, b3, b4,b5)=(0, 0, 0, 0, 1, 1) is mapped onto (I, Q)=(5×k, 5×k), (b0, b1, b2,b3, b4, b5)=(0, 0, 0, 1, 0, 0) is mapped onto (I, Q)=(7×k, 1×k), . . . ,(b0, b1, b2, b3, b4, b5)=(1, 1, 1, 1, 1, 0) is mapped onto (I, Q)=(−3×k,−1×k), and (b0, b1, b2, b3, b4, b5)=(1, 1, 1, 1, 1, 1) is mapped onto(I, Q)=(−3×k, −3×k). Note that b0 through b5 shown on the top right ofFIG. 86 shows the bits and the arrangement of the numbers shown on theIQ plane.

In FIG. 84, the power changer 8401B sets such that u=u₁₆ when themodulation scheme for s2 is 16QAM, and sets such that u=u₆₄ when themodulation scheme for s2 is 64QAM. In this case, due to the relationshipbetween minimum Euclidian distances, by setting such that u₁₆<u₆₄,excellent data reception quality is obtained in the reception devicewhen the modulation scheme for s2 is either 16QAM or 64QAM.

Note that, in the above description, the “modulation scheme for s1 isfixed to QPSK”. It is also considered that the modulation scheme for s2is fixed to QPSK. In this case, power change is assumed to be notperformed for the fixed modulation scheme (here, QPSK), and to beperformed for a plurality of modulation schemes that can be set (here,16QAM and 64QAM). That is to say, in this case, the transmission devicedoes not have the structure shown in FIG. 84, but has a structure inwhich the power changer 8401B is eliminated from the structure in FIG.84 and a power changer is provided to a s1(t)-side. When the fixedmodulation scheme (here, QPSK) is set to s2, the following formula 86 issatisfied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 86} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ue}^{j\; 0} & 0 \\0 & e^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}u & 0 \\0 & 1\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu} 86} \right)\end{matrix}$

When the modulation scheme for s2 is fixed to QPSK and the modulationscheme for s1 is changed from 16QAM to 64QAM (is set to either 16QAM or64QAM), the relationship u₁₆<u₆₄ should be satisfied (note that amultiplied value for power change in 16QAM is u₁₆, a multiplied valuefor power change in 64QAM is u₆₄, and power change is not performed inQPSK).

Also, when a set of the modulation scheme for s1 and the modulationscheme for s2 can be set to any one of a set of QPSK and 16QAM, a set of16QAM and QPSK, a set of QPSK and 64QAM and a set of 64QAM and QPSK, therelationship u₁₆<u₆₄ should be satisfied.

The following describes a case where the above-mentioned description isgeneralized.

Let the modulation scheme for s1 be fixed to a modulation scheme C inwhich the number of signal points in the IQ plane is c. Also, let themodulation scheme for s2 be set to either a modulation scheme A in whichthe number of signal points in the IQ plane is a or a modulation schemeB in which the number of signal points in the IQ plane is b (a>b>c)(however, let the average power (average value) for s2 in the modulationscheme A be equal to the average power (average value) for s2 in themodulation scheme B).

In this case, a value for power change set when the modulation scheme Ais set to the modulation scheme for s2 is u_(a). Also, a value for powerchange set when the modulation scheme B is set to the modulation schemefor s2 is u_(b). In this case, when the relationship u_(b)<u_(a) issatisfied, excellent data reception quality is obtained in the receptiondevice.

Power change is assumed to be not performed for the fixed modulationscheme (here, modulation scheme C), and to be performed for a pluralityof modulation schemes that can be set (here, modulation schemes A andB). When the modulation scheme for s2 is fixed to the modulation schemeC and the modulation scheme for s1 is changed from the modulation schemeA to the modulation scheme B (is set to either the modulation schemes Aor B), the relationship u_(b)<u_(a) should be satisfied. Also, when aset of the modulation scheme for s1 and the modulation scheme for s2 canbe set to any one of a set of the modulation scheme C and the modulationscheme A, a set of the modulation scheme A and the modulation scheme C,a set of the modulation scheme C and the modulation scheme B and a setof the modulation scheme B and the modulation scheme C, the relationshipu_(b)<u_(a) should be satisfied.

Example 2

The following describes an example of the operation different from thatdescribed in Example 1, using FIG. 84. Let s1(t) be the (mapped)baseband signal for the modulation scheme 64QAM. The mapping scheme fors1(t) is as shown in FIG. 86, and k is as represented by formula 85.Also, let s2(t) be the (mapped) baseband signal for the modulationscheme 16QAM. The mapping scheme for s2(t) is as shown in FIG. 80, and gis as represented by formula 79. Note that t is time. In the presentembodiment, description is made taking the time domain as an example.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 16QAM by u.Let u be a real number, and u<1.0. Letting the precoding matrix used inthe scheme for regularly performing phase change on the modulated signalafter precoding be F and the phase changing value used for regularlyperforming phase change be y(t) (y(t) may be imaginary number having theabsolute value of 1, i.e. ej^(θ(t)), formula 82 is satisfied.

Therefore, a ratio of the average power for 64QAM to the average powerfor 16QAM is set to 1:u². With this structure, the reception device isin a reception condition as shown in FIG. 83. Therefore, data receptionquality is improved in the reception device.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the value u for power change is set based onthe control signal (8400). The following describes setting of the valueu for power change based on the control signal (8400) in order toimprove data reception quality in the reception device in detail.

Example 2-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction codes used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-1 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected block lengthindicated by the control signal (8400). Here, a value for power changeset according to a block length X is referred to as u_(LX)

For example, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000). Inthis case, for example, by setting u_(L1000), u_(L1500) and u_(L3000) soas to be different from one another, a high error correction capabilitycan be achieved for each code length. Depending on the set code length,however, the effect might not be obtained even if the value for powerchange is changed. In such a case, even when the code length is changed,it is unnecessary to change the value for power change (for example,u_(L1000)=u_(L1500) may be satisfied. What is important is that two ormore values exist in u_(L1000), u_(L1500) and u_(L3000)).

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more code lengths that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the code length is set, and performspower change.

Example 2-2

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a coding rate for the error correctioncodes used to generate s1 and s2 when the transmission device supports aplurality of coding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-2 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected coding rateindicated by the control signal (8400). Here, a value for power changeset according to a coding rate _(rx) is referred to as u_(rx).

For example, when r1 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r1). When r2 is selected asthe coding rate, the power changer (8401B) sets a value for power changeto u_(r2). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3). In this case, forexample, by setting u_(r1), u_(r2) and u_(r3) so as to be different fromone another, a high error correction capability can be achieved for eachcoding rate. Depending on the set coding rate, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the coding rate is changed, it is unnecessary to changethe value for power change (for example, u_(r1)=u_(r2) may be satisfied.What is important is that two or more values exist in u_(r1), u_(r2) andu_(r3)). Note that, as examples of r1, r2 and r3 described above, codingrates 1/2, 2/3 and 3/4 are considered when the error correction code isthe LDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more coding rates that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the coding rate is set, and performspower change.

Example 2-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto 64QAM and the modulation scheme for s2 is changed from 16QAM to QPSKby the control signal (or can be set to either 16QAM or QPSK) isconsidered. In a case where the modulation scheme for s1 is 64QAM, themapping scheme for s1(t) is as shown in FIG. 86, and k is represented byformula 85 in FIG. 86. In a case where the modulation scheme for s2 is16QAM, the mapping scheme for s2(t) is as shown in FIG. 80, and g isrepresented by formula 79 in FIG. 80. Also, in a case where themodulation scheme for s2(t) is QPSK, the mapping scheme for s2(t) is asshown in FIG. 81, and h is represented by formula 78 in FIG. 81.

By performing mapping in this way, the average power in 16QAM becomesequal to the average power (average value) in QPSK.

In FIG. 84, the power changer 8401B sets such that u=u₁₆ when themodulation scheme for s2 is 16QAM, and sets such that u=u₄ when themodulation scheme for s2 is QPSK. In this case, due to the relationshipbetween minimum Euclidian distances, by setting such that u₄<u₁₆,excellent data reception quality is obtained in the reception devicewhen the modulation scheme for s2 is either 16QAM or QPSK.

Note that, in the above description, the modulation scheme for s1 isfixed to 64QAM. When the modulation scheme for s2 is fixed to 64QAM andthe modulation scheme for s1 is changed from 16QAM to QPSK (is set toeither 16QAM or QPSK), the relationship u₄<u₁₆ should be satisfied (thesame considerations should be made as the example 1-3) (note that amultiplied value for power change in 16QAM is u₁₆, a multiplied valuefor power change in QPSK is u₄, and power change is not performed in64QAM). Also, when a set of the modulation scheme for s1 and themodulation scheme for s2 can be set to any one of a set of 64QAM and16QAM, a set of 16QAM and 64QAM, a set of 64QAM and QPSK and a set ofQPSK and 64QAM, the relationship u₄<u₁₆ should be satisfied.

The following describes a case where the above-mentioned description isgeneralized.

Let the modulation scheme for s1 be fixed to a modulation scheme C inwhich the number of signal points in the IQ plane is c. Also, let themodulation scheme for s2 be set to either a modulation scheme A in whichthe number of signal points in the IQ plane is a or a modulation schemeB in which the number of signal points in the IQ plane is b (c>b>a)(however, let the average power (average value) for s2 in the modulationscheme A be equal to the average power (average value) for s2 in themodulation scheme B).

In this case, a value for power change set when the modulation scheme Ais set to the modulation scheme for s2 is u_(a). Also, a value for powerchange set when the modulation scheme B is set to the modulation schemefor s2 is u_(b). In this case, when the relationship u_(a)<u_(b) issatisfied, excellent data reception quality is obtained in the receptiondevice.

Power change is assumed to be not performed for the fixed modulationscheme (here, modulation scheme C), and to be performed for a pluralityof modulation schemes that can be set (here, modulation schemes A andB). When the modulation scheme for s2 is fixed to the modulation schemeC and the modulation scheme for s1 is changed from the modulation schemeA to the modulation scheme B (is set to either the modulation schemes Aor B), the relationship u_(a)<u_(b) should be satisfied. Also, when aset of the modulation scheme for s1 and the modulation scheme for s2 canbe set to any one of a set of the modulation scheme C and the modulationscheme A, a set of the modulation scheme A and the modulation scheme C,a set of the modulation scheme C and the modulation scheme B and a setof the modulation scheme B and the modulation scheme C, the relationshipu_(a)<u_(b) should be satisfied.

Example 3

The following describes an example of the operation different from thatdescribed in Example 1, using FIG. 84. Let s1(t) be the (mapped)baseband signal for the modulation scheme 16QAM. The mapping scheme fors1(t) is as shown in FIG. 80, and g is as represented by formula 79. Lets2(t) be the (mapped) baseband signal for the modulation scheme 64QAM.The mapping scheme for s2(t) is as shown in FIG. 86, and k is asrepresented by formula 85. Note that t is time. In the presentembodiment, description is made taking the time domain as an example.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 64QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 64QAM by u.Let u be a real number, and u>1.0. Letting the precoding matrix used inthe scheme for regularly performing phase change on the modulated signalafter precoding be F and the phase changing value used for regularlyperforming phase change be y(t) (y(t) may be imaginary number having theabsolute value of 1, i.e. ej^(θ(t)), formula 82 is satisfied.

Therefore, a ratio of the average power for 16QAM to the average powerfor 64QAM is set to 1:u². With this structure, the reception device isin a reception condition as shown in FIG. 83. Therefore, data receptionquality is improved in the reception device.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the value u for power change is set based onthe control signal (8400). The following describes setting of the valueu for power change based on the control signal (8400) in order toimprove data reception quality in the reception device in detail.

Example 3-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction codes used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-1 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected block lengthindicated by the control signal (8400). Here, a value for power changeset according to a block length X is referred to as u_(LX)

For example, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000). Inthis case, for example, by setting u_(L1000), u_(L1500) and u_(L3000) soas to be different from one another, a high error correction capabilitycan be achieved for each code length. Depending on the set code length,however, the effect might not be obtained even if the value for powerchange is changed. In such a case, even when the code length is changed,it is unnecessary to change the value for power change (for example,u_(L1000)=u_(L1500) may be satisfied. What is important is that two ormore values exist in u_(L1000), u_(L1500) and u_(L3000)).

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more code lengths that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the code length is set, and performspower change.

Example 3-2

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a coding rate for the error correctioncodes used to generate s1 and s2 when the transmission device supports aplurality of coding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401B) sets the value u for power change according to the controlsignal (8400).

The example 1-2 is characterized in that the power changer (8401B) setsthe value u for power change according to the selected coding rateindicated by the control signal (8400). Here, a value for power changeset according to a coding rate rx is referred to as u_(rx).

For example, when r1 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r1). When r2 is selected asthe coding rate, the power changer (8401B) sets a value for power changeto u_(r2). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3). In this case, forexample, by setting u_(r1), u_(r2) and u_(r3) so as to be different fromone another, a high error correction capability can be achieved for eachcoding rate. Depending on the set coding rate, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the coding rate is changed, it is unnecessary to changethe value for power change (for example, u_(r1)=u_(r2) may be satisfied.What is important is that two or more values exist in u_(r1), u_(r2) andu_(r3)).

Note that, as examples of r1, r2 and r3 described above, coding rates1/2, 2/3 and 3/4 are considered when the error correction code is theLDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Theimportant point is that two or more values for power change exist whenthere are two or more coding rates that can be set, and the transmissiondevice selects any of the values for power change from among the two ormore values for power change when the coding rate is set, and performspower change.

Example 3-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto 16QAM and the modulation scheme for s2 is changed from 64QAM to QPSKby the control signal (or can be set to either 64QAM or QPSK) isconsidered. In a case where the modulation scheme for s1 is 16QAM, themapping scheme for s2(t) is as shown in FIG. 80, and g is represented byformula 79 in FIG. 80. In a case where the modulation scheme for s2 is64QAM, the mapping scheme for s1(t) is as shown in FIG. 86, and k isrepresented by formula 85 in FIG. 86. Also, in a case where themodulation scheme for s2(t) is QPSK, the mapping scheme for s2(t) is asshown in FIG. 81, and h is represented by formula 78 in FIG. 81.

By performing mapping in this way, the average power in 16QAM becomesequal to the average power in QPSK.

In FIG. 84, the power changer 8401B sets such that u=u₆₄ when themodulation scheme for s2 is 64QAM, and sets such that u=u₄ when themodulation scheme for s2 is QPSK. In this case, due to the relationshipbetween minimum Euclidian distances, by setting such that u₄<u₆₄,excellent data reception quality is obtained in the reception devicewhen the modulation scheme for s2 is either 16QAM or 64QAM.

Note that, in the above description, the modulation scheme for s1 isfixed to 16QAM. When the modulation scheme for s2 is fixed to 16QAM andthe modulation scheme for s1 is changed from 64QAM to QPSK (is set toeither 64QAM or QPSK), the relationship u₄<u₆₄ should be satisfied (thesame considerations should be made as the example 1-3) (note that amultiplied value for power change in 64QAM is u₆₄, a multiplied valuefor power change in QPSK is u₄, and power change is not performed in16QAM). Also, when a set of the modulation scheme for s1 and themodulation scheme for s2 can be set to any one of a set of 16QAM and64QAM, a set of 64QAM and 16QAM, a set of 16QAM and QPSK and a set ofQPSK and 16QAM, the relationship u₄<u₆₄ should be satisfied.

The following describes a case where the above-mentioned description isgeneralized.

Let the modulation scheme for s1 be fixed to a modulation scheme C inwhich the number of signal points in the IQ plane is c. Also, let themodulation scheme for s2 be set to either a modulation scheme A in whichthe number of signal points in the IQ plane is a or a modulation schemeB in which the number of signal points in the IQ plane is b (c>b>a)(however, let the average power (average value) for s2 in the modulationscheme A be equal to the average power (average value) for s2 in themodulation scheme B).

In this case, a value for power change set when the modulation scheme Ais set to the modulation scheme for s2 is u_(a). Also, a value for powerchange set when the modulation scheme B is set to the modulation schemefor s2 is u_(b). In this case, when the relationship u_(a)<u_(b) issatisfied, excellent data reception quality is obtained in the receptiondevice.

Power change is assumed to be not performed for the fixed modulationscheme (here, modulation scheme C), and to be performed for a pluralityof modulation schemes that can be set (here, modulation schemes A andB). When the modulation scheme for s2 is fixed to the modulation schemeC and the modulation scheme for s1 is changed from the modulation schemeA to the modulation scheme B (is set to either the modulation schemes Aor B), the relationship u_(a)<u_(b) should be satisfied. Also, when aset of the modulation scheme for s1 and the modulation scheme for s2 canbe set to any one of a set of the modulation scheme C and the modulationscheme A, a set of the modulation scheme A and the modulation scheme C,a set of the modulation scheme C and the modulation scheme B and a setof the modulation scheme B and the modulation scheme C, the relationshipu_(a)<u_(b) should be satisfied.

Example 4

The case where power change is performed for one of the modulationschemes for s1 and s2 has been described above. The following describesa case where power change is performed for both of the modulationschemes for s1 and s2.

An example of the operation is described using FIG. 85. Let s1(t) be the(mapped) baseband signal for the modulation scheme QPSK. The mappingscheme for s1(t) is as shown in FIG. 81, and h is as represented byformula 78. Also, let s2(t) be the (mapped) baseband signal for themodulation scheme 16QAM. The mapping scheme for s2(t) is as shown inFIG. 80, and g is as represented by formula 79. Note that t is time. Inthe present embodiment, description is made taking the time domain as anexample.

The power changer (8401A) receives a (mapped) baseband signal 307A forthe modulation scheme QPSK and the control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be v, the power changer outputs a signal (8402A) obtained by multiplyingthe (mapped) baseband signal 307A for the modulation scheme QPSK by v.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 16QAM by u.Then, let u=v×w (w>1.0).

Letting the precoding matrix used in the scheme for regularly performingphase change on the modulated signal after precoding be F and the phasechanging value used for regularly performing phase change be y(t) (y(t)may be imaginary number having the absolute value of 1, i.e. ej^(θ(t)),formula 87 shown next is satisfied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 87} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & {v \times w}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu} 87} \right)\end{matrix}$

Therefore, a ratio of the average power for QPSK to the average powerfor 16QAM is set to v²:u²=v²:v²×w²=1:w². With this structure, thereception device is in a reception condition as shown in FIG. 83.Therefore, data reception quality is improved in the reception device.

Note that, in view of formula 83 and formula 84, effective examples ofthe ratio of the average power for QPSK to the average power for 16QAMare considered to be v²:u²=v²:v²×w²=1:w²=1:5 or v²:u²=v²:v²×w²=1:w²=1:2.The ratio, however, is set appropriately according to conditionsrequired as a system.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the values v and u for power change are setbased on the control signal (8400). The following describes setting ofthe values v and u for power change based on the control signal (8400)in order to improve data reception quality in the reception device indetail.

Example 4-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction codes used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected block length indicated by the control signal(8400). Here, values for power change set according to the block lengthX are referred to as v_(LX) and u_(LX).

For example, when 1000 is selected as the block length, the powerchanger (8401A) sets a value for power change to V_(L1000). When 1500 isselected as the block length, the power changer (8401A) sets a value forpower change to v_(L1500). When 3000 is selected as the block length,the power changer (8401A) sets a value for power change to v_(L3000).

On the other hand, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000).

In this case, for example, by setting v_(L1000), v_(L1500) and v_(L3000)so as to be different from one another, a high error correctioncapability can be achieved for each code length. Similarly, by settingu_(L1000), u_(L1500) and u_(L3000) so as to be different from oneanother, a high error correction capability can be achieved for eachcode length. Depending on the set code length, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the code length is changed, it is unnecessary to changethe value for power change (for example, u_(L1000)=u_(L1500) may besatisfied, and V_(L1000)=v_(L1500) may be satisfied. What is importantis that two or more values exist in a set of v_(L1000), v_(L1500) andv_(L3000), and that two or more values exist in a set of u_(L1000),u_(L1500) and u_(L3000)). Note that, as described above, v_(LX) andu_(LX) are set so as to satisfy the ratio of the average power 1:w².

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u_(LX) for power change existwhen there are two or more code lengths that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(LX) for power change when the codelength is set, and performs power change. Another important point isthat two or more values v_(LX) for power change exist when there are twoor more code lengths that can be set, and the transmission deviceselects any of the values for power change from among the two or morevalues v_(LX) for power change when the code length is set, and performspower change.

Example 4-2

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a coding rate for the error correctioncodes used to generate s1 and s2 when the transmission device supports aplurality of coding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401A) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected coding rate indicated by the control signal(8400). Here, values for power change set according to the coding raterx are referred to as v_(rx) and u_(rx).

For example, when r1 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r1). When r2 is selected asthe coding rate, the power changer (8401A) sets a value for power changeto v_(r2). When r3 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r3).

Also, when r1 is selected as the coding rate, the power changer (8401B)sets a value for power change to u_(r1). When r2 is selected as thecoding rate, the power changer (8401B) sets a value for power change tou_(r3). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3).

In this case, for example, by setting v_(r1), v_(r2) and v_(r3) so as tobe different from one another, a high error correction capability can beachieved for each code length. Similarly, by setting u_(r1), u_(r2) andu_(r3) so as to be different from one another, a high error correctioncapability can be achieved for each coding rate. Depending on the setcoding rate, however, the effect might not be obtained even if the valuefor power change is changed. In such a case, even when the coding rateis changed, it is unnecessary to change the value for power change (forexample, v_(r1)=v_(r2) may be satisfied, and u_(r1)=u_(r3) may besatisfied. What is important is that two or more values exist in a setof v_(r1), v_(r2) and v_(r3), and that two or more values exist in a setof u_(r1), u_(r2) and u_(r3)). Note that, as described above, v_(rX) andu_(rX) are set so as to satisfy the ratio of the average power 1:w².

Also, note that, as examples of r1, r2 and r3 described above, codingrates 1/2, 2/3 and 3/4 are considered when the error correction code isthe LDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u, for power change existwhen there are two or more coding rates that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(rx) for power change when the codingrate is set, and performs power change. Another important point is thattwo or more values v_(rX) for power change exist when there are two ormore coding rates that can be set, and the transmission device selectsany of the values for power change from among the two or more valuesv_(rX) for power change when the coding rate is set, and performs powerchange.

Example 4-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto QPSK and the modulation scheme for s2 is changed from 16QAM to 64QAMby the control signal (or can be set to either 16QAM or 64QAM) isconsidered. In a case where the modulation scheme for s1 is QPSK, themapping scheme for s1(t) is as shown in FIG. 81, and h is represented byformula 78 in FIG. 81. In a case where the modulation scheme for s2 is16QAM, the mapping scheme for s2(t) is as shown in FIG. 80, and g isrepresented by formula 79 in FIG. 80. Also, in a case where themodulation scheme for s2(t) is 64QAM, the mapping scheme for s2(t) is asshown in FIG. 86, and k is represented by formula 85 in FIG. 86.

In FIG. 85, when the modulation scheme for s1 is QPSK and the modulationscheme for s2 is 16QAM, assume that v=α and u=α×w₁₆. In this case, theratio between the average power of QPSK and the average power of 16QAMis v²:u²=α²:α₂×w₁₆ ²=1:w₁₆ ².

In FIG. 85, when the modulation scheme for s1 is QPSK and the modulationscheme for s2 is 64QAM, assume that v=β and u=β×w₆₄. In this case, theratio between the average power of QPSK and the average power of 64QAMis v:u=β²:β²×w₆₄ ²=1:w₆₄ ². In this case, according to the minimumEuclidean distance relationship, the reception device achieves high datareception quality when 1.0<w₁₆<w₆₄, regardless of whether the modulationscheme for s2 is 16QAM or 64QAM.

Note that although “the modulation scheme for s1 is fixed to QPSK” inthe description above, it is possible that “the modulation scheme for s2is fixed to QPSK”. In this case, power change is assumed to be notperformed for the fixed modulation scheme (here, QPSK), and to beperformed for a plurality of modulation schemes that can be set (here,16QAM and 64QAM). When the fixed modulation scheme (here, QPSK) is setto s2, the following formula 88 is satisfied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 88} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ue}^{j\; 0} & 0 \\0 & {ve}^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}u & 0 \\0 & v\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{v \times w} & 0 \\0 & v\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu} 88} \right)\end{matrix}$

Given that, even when “the modulation scheme for s2 is fixed to QPSK andthe modulation scheme for s1 is changed from 16QAM to 64QAM (set toeither 16QAM or 64QAM)”, 1.0<w₁₆<w₆₄ should be fulfilled. (Note that thevalue used for the multiplication for the power change in the case of16QAM is u=α×w₆, the value used for the multiplication for the powerchange in the case of 64QAM is u=β×w₆₄, the value used for the powerchange in the case of QPSK is v=α when the selectable modulation schemeis 16QAM and v=β when the selectable modulation scheme is 64QAM.) Also,when the set of (the modulation scheme for s1, the modulation scheme fors2) is selectable from the sets of (QPSK, 16QAM), (16QAM, QPSK), (QPSK,64QAM) and (64QAM, QPSK), 1.0<w₁₆<w₆₄ should be fulfilled.

The following describes a case where the above-mentioned description isgeneralized.

For generalization, assume that the modulation scheme for s1 is fixed toa modulation scheme C with which the number of signal points in the IQplane is c. Also assume that the modulation scheme for s2 is selectablefrom a modulation scheme A with which the number of signal points in theIQ plane is a and a modulation scheme B with which the number of signalpoints in the IQ plane is b (a>b>c). In this case, when the modulationscheme for s2 is set to the modulation scheme A, assume that ratiobetween the average power of the modulation scheme for s1, which is themodulation scheme C, and the average power of the modulation scheme fors2, which is the modulation scheme A, is 1:w_(a) ². Also, when themodulation scheme for s2 is set to the modulation scheme B, assume thatratio between the average power of the modulation scheme for s1, whichis the modulation scheme C, and the average power of the modulationscheme for s2, which is the modulation scheme B, is 1:w_(b) ². If thisis the case, the reception device achieves a high data reception qualitywhen w_(b)<W_(a) is fulfilled.

Note that although “the modulation scheme for s1 is fixed to C” in thedescription above, even when “the modulation scheme for s2 is fixed tothe modulation scheme C and the modulation scheme for s1 is changed fromthe modulation scheme A to the modulation scheme B (set to either themodulation scheme A or the modulation scheme B), the average powersshould fulfill w_(b)<w_(a). (If this is the case, as with thedescription above, when the average power of the modulation scheme C is1, the average power of the modulation scheme A is w_(a) ², and theaverage power of the modulation scheme B is w_(b) ².) Also, when the setof (the modulation scheme for s1, the modulation scheme for s2) isselectable from the sets of (the modulation scheme C, the modulationscheme A), (the modulation scheme A, the modulation scheme C), (themodulation scheme C, the modulation scheme B) and (the modulation schemeB, the modulation scheme C), the average powers should fulfillw_(b)<W_(a).

Example 5

The following describes an example of the operation different from thatdescribed in Example 4, using FIG. 85. Let s1(t) be the (mapped)baseband signal for the modulation scheme 64QAM. The mapping scheme fors1(t) is as shown in FIG. 86, and k is as represented by formula 85.Also, let s2(t) be the (mapped) baseband signal for the modulationscheme 16QAM. The mapping scheme for s2(t) is as shown in FIG. 80, and gis as represented by formula 79. Note that t is time. In the presentembodiment, description is made taking the time domain as an example.

The power changer (8401A) receives a (mapped) baseband signal 307A forthe modulation scheme 64QAM and the control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be v, the power changer outputs a signal (8402A) obtained by multiplyingthe (mapped) baseband signal 307A for the modulation scheme 64QAM by v.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 16QAM by u.Then, let u=v×w (w<1.0).

Letting the precoding matrix used in the scheme for regularly performingphase change on the modulated signal after precoding be F and the phasechanging value used for regularly performing phase change be y(t) (y(t)may be imaginary number having the absolute value of 1, i.e. ej^(θ(t)),formula 87 shown above is satisfied.

Therefore, a ratio of the average power for 64QAM to the average powerfor 16QAM is set to v²:u²=v²:v²×w²=1:w². With this structure, thereception device is in a reception condition as shown in FIG. 83.Therefore, data reception quality is improved in the reception device.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the values v and u for power change are setbased on the control signal (8400). The following describes setting ofthe values v and u for power change based on the control signal (8400)in order to improve data reception quality in the reception device indetail.

Example 5-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction codes used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected block length indicated by the control signal(8400). Here, values for power change set according to the block lengthX are referred to as v_(LX) and u_(LX).

For example, when 1000 is selected as the block length, the powerchanger (8401A) sets a value for power change to V_(L1000). When 1500 isselected as the block length, the power changer (8401A) sets a value forpower change to v_(L1500). When 3000 is selected as the block length,the power changer (8401A) sets a value for power change to v_(L3000).

On the other hand, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000).

In this case, for example, by setting v_(L1000), v_(L1500) and v_(L3000)so as to be different from one another, a high error correctioncapability can be achieved for each code length. Similarly, by settingu_(L1000), u_(L1500) and u_(L3000) so as to be different from oneanother, a high error correction capability can be achieved for eachcode length. Depending on the set code length, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the code length is changed, it is unnecessary to changethe value for power change (for example, u_(L1000)=u_(L1500) may besatisfied, and V_(L1000)=v_(L1500) may be satisfied. What is importantis that two or more values exist in a set of v_(L1000), v_(L1500) andv_(L3000), and that two or more values exist in a set of u_(L1000),u_(L1500) and u_(L3000)). Note that, as described above, v_(LX) andu_(LX) are set so as to satisfy the ratio of the average power 1:w².

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u_(LX) for power change existwhen there are two or more code lengths that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(LX) for power change when the codelength is set, and performs power change. Another important point isthat two or more values v_(LX) for power change exist when there are twoor more code lengths that can be set, and the transmission deviceselects any of the values for power change from among the two or morevalues v_(LX) for power change when the code length is set, and performspower change.

Example 5-2

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a coding rate for the error correctioncodes used to generate s1 and s2 when the transmission device supports aplurality of coding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401A) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected coding rate indicated by the control signal(8400). Here, values for power change set according to the coding raterx are referred to as v_(rx) and u_(rx).

For example, when r1 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r1). When r2 is selected asthe coding rate, the power changer (8401A) sets a value for power changeto v_(r2). When r3 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r3).

Also, when r1 is selected as the coding rate, the power changer (8401B)sets a value for power change to u_(r1). When r2 is selected as thecoding rate, the power changer (8401B) sets a value for power change tou_(r3). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3).

In this case, for example, by setting v_(r1), v_(r2) and v_(r3) so as tobe different from one another, a high error correction capability can beachieved for each code length. Similarly, by setting u_(r1), u_(r2) andu_(r3) so as to be different from one another, a high error correctioncapability can be achieved for each coding rate. Depending on the setcoding rate, however, the effect might not be obtained even if the valuefor power change is changed. In such a case, even when the coding rateis changed, it is unnecessary to change the value for power change (forexample, v_(r1)=v_(r2) may be satisfied, and u_(r1)=u_(r3) may besatisfied. What is important is that two or more values exist in a setof v_(r1), v_(r2) and v_(r3), and that two or more values exist in a setof u_(r1), u_(r2) and u_(r3)). Note that, as described above, v_(rX) andu_(rX) are set so as to satisfy the ratio of the average power 1:w².

Also, note that, as examples of r1, r2 and r3 described above, codingrates 1/2, 2/3 and 3/4 are considered when the error correction code isthe LDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u, for power change existwhen there are two or more coding rates that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(rx) for power change when the codingrate is set, and performs power change. Another important point is thattwo or more values v_(rX) for power change exist when there are two ormore coding rates that can be set, and the transmission device selectsany of the values for power change from among the two or more valuesv_(rX) for power change when the coding rate is set, and performs powerchange.

Example 5-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto 64QAM and the modulation scheme for s2 is changed from 16QAM to QPSKby the control signal (or can be set to either 16QAM or QPSK) isconsidered. In a case where the modulation scheme for s1 is 64QAM, themapping scheme for s1(t) is as shown in FIG. 86, and k is represented byformula 85 in FIG. 86. In a case where the modulation scheme for s2 is16QAM, the mapping scheme for s2(t) is as shown in FIG. 80, and g isrepresented by formula 79 in FIG. 80. Also, in a case where themodulation scheme for s2(t) is QPSK, the mapping scheme for s2(t) is asshown in FIG. 81, and h is represented by formula 78 in FIG. 81.

In FIG. 85, when the modulation scheme for s1 is 64QAM and themodulation scheme for s2 is 16QAM, assume that v=α and u=α×w₁₆. In thiscase, the ratio between the average power of 64QAM and the average powerof 16QAM is v²:u²=α²:α²×w₁₆ ²=1:w₁₆ ².

In FIG. 85, when the modulation scheme for s1 is 64QAM and themodulation scheme for s2 is QPSK, assume that v=β and u=β×w₄. In thiscase, the ratio between the average power of 64QAM and the average powerof QPSK is v²:u²=β²:β²×w₄ ²=1:w₄ ². In this case, according to theminimum Euclidean distance relationship, the reception device achieves ahigh data reception quality when w₄<W₁₆<1.0, regardless of whether themodulation scheme for s2 is 16QAM or QPSK.

Note that although “the modulation scheme for s1 is fixed to 64QAM” inthe description above, it is possible that “the modulation scheme for s2is fixed to 64QAM and the modulation scheme for s1 is changed from 16QAMto QPSK (set to either 16QAM or QPSK)”, w₄<w₁₆<1.0 should be fulfilled.(The same as described in Example 4-3). (Note that the value used forthe multiplication for the power change in the case of 16QAM is u=α×w₁₆,the value used for the multiplication for the power change in the caseof QPSK is u=β×w₄, the value used for the power change in the case of64QAM is v=α when the selectable modulation scheme is 16QAM and v=β whenthe selectable modulation scheme is QPSK). Also, when the set of (themodulation scheme for s1, the modulation scheme for s2) is selectablefrom the sets of (64QAM, 16QAM), (16QAM, 64QAM), (64QAM, QPSK) and(QPSK, 64QAM), w₄<w₁₆<1.0 should be fulfilled.

The following describes a case where the above-mentioned description isgeneralized.

For generalization, assume that the modulation scheme for s1 is fixed toa modulation scheme C with which the number of signal points in the IQplane is c. Also assume that the modulation scheme for s2 is selectablefrom a modulation scheme A with which the number of signal points in theIQ plane is a and a modulation scheme B with which the number of signalpoints in the IQ plane is b (c>b>a). In this case, when the modulationscheme for s2 is set to the modulation scheme A, assume that ratiobetween the average power of the modulation scheme for s1, which is themodulation scheme C, and the average power of the modulation scheme fors2, which is the modulation scheme A, is 1:w_(a) ². Also, when themodulation scheme for s2 is set to the modulation scheme B, assume thatratio between the average power of the modulation scheme for s1, whichis the modulation scheme C, and the average power of the modulationscheme for s2, which is the modulation scheme B, is 1:w_(b) ². If thisis the case, the reception device achieves a high data reception qualitywhen w_(a)<w_(b) is fulfilled.

Note that although “the modulation scheme for s1 is fixed to C” in thedescription above, even when “the modulation scheme for s2 is fixed tothe modulation scheme C and the modulation scheme for s1 is changed fromthe modulation scheme A to the modulation scheme B (set to either themodulation scheme A or the modulation scheme B), the average powersshould fulfill w_(a)<w_(b). (If this is the case, as with thedescription above, when the average power of the modulation scheme is C,the average power of the modulation scheme A is w_(a) ², and the averagepower of the modulation scheme B is w_(b) ².) Also, when the set of (themodulation scheme for s1, the modulation scheme for s2) is selectablefrom the sets of (the modulation scheme C, the modulation scheme A),(the modulation scheme A, the modulation scheme C), (the modulationscheme C, the modulation scheme B) and (the modulation scheme B, themodulation scheme C), the average powers should fulfill w_(a)<W_(b).

Example 6

The following describes an example of the operation different from thatdescribed in Example 4, using FIG. 85. Let s1(t) be the (mapped)baseband signal for the modulation scheme 16QAM. The mapping scheme fors1(t) is as shown in FIG. 86, and g is as represented by formula 79. Lets2(t) be the (mapped) baseband signal for the modulation scheme 64QAM.The mapping scheme for s2(t) is as shown in FIG. 86, and k is asrepresented by formula 85. Note that t is time. In the presentembodiment, description is made taking the time domain as an example.

The power changer (8401A) receives a (mapped) baseband signal 307A forthe modulation scheme 16QAM and the control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be v, the power changer outputs a signal (8402A) obtained by multiplyingthe (mapped) baseband signal 307A for the modulation scheme 16QAM by v.

The power changer (8401B) receives a (mapped) baseband signal 307B forthe modulation scheme 64QAM and a control signal (8400) as input.Letting a value for power change set based on the control signal (8400)be u, the power changer outputs a signal (8402B) obtained by multiplyingthe (mapped) baseband signal 307B for the modulation scheme 64QAM by u.Then, let u=v×w (w<1.0).

Letting the precoding matrix used in the scheme for regularly performingphase change on the modulated signal after precoding be F and the phasechanging value used for regularly performing phase change be y(t) (y(t)may be imaginary number having the absolute value of 1, i.e. ej^(θ(t)),formula 87 shown above is satisfied.

Therefore, a ratio of the average power for 64QAM to the average powerfor 16QAM is set to v²:u²=v²:v²×w²=1:w². With this structure, thereception device is in a reception condition as shown in FIG. 83.Therefore, data reception quality is improved in the reception device.

In the conventional technology, transmission power control is generallyperformed based on feedback information from a communication partner.The present invention is characterized in that the transmission power iscontrolled regardless of the feedback information from the communicationpartner in the present embodiment. Detailed description is made on thispoint.

The above describes that the values v and u for power change are setbased on the control signal (8400). The following describes setting ofthe values v and u for power change based on the control signal (8400)in order to improve data reception quality in the reception device indetail.

Example 6-1

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a block length (the number of bitsconstituting one coding (encoded) block, and is also referred to as thecode length) for the error correction codes used to generate s1 and s2when the transmission device supports a plurality of block lengths forthe error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of block lengths are supported. Encoded data for which errorcorrection codes whose block length is selected from among the pluralityof supported block lengths has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected blocklength for the error correction codes described above. The power changer(8401B) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected block length indicated by the control signal(8400). Here, values for power change set according to the block lengthX are referred to as v_(LX) and u_(LX).

For example, when 1000 is selected as the block length, the powerchanger (8401A) sets a value for power change to v_(L1000). When 1500 isselected as the block length, the power changer (8401A) sets a value forpower change to v_(L1500). When 3000 is selected as the block length,the power changer (8401A) sets a value for power change to v_(L3000).

On the other hand, when 1000 is selected as the block length, the powerchanger (8401B) sets a value for power change to u_(L1000). When 1500 isselected as the block length, the power changer (8401B) sets a value forpower change to u_(L1500). When 3000 is selected as the block length,the power changer (8401B) sets a value for power change to u_(L3000).

In this case, for example, by setting v_(L1000), v_(L1500) and v_(L3000)so as to be different from one another, a high error correctioncapability can be achieved for each code length. Similarly, by settingu_(L1000), u_(L1500) and u_(L3000) so as to be different from oneanother, a high error correction capability can be achieved for eachcode length. Depending on the set code length, however, the effect mightnot be obtained even if the value for power change is changed. In such acase, even when the code length is changed, it is unnecessary to changethe value for power change (for example, u_(L1000)=u_(L1500) may besatisfied, and v_(L1000)=v_(L1500) may be satisfied. What is importantis that two or more values exist in a set of v_(L1000), v_(L1500) andv_(L3000), and that two or more values exist in a set of u_(L1000),u_(L1500) and u_(L3000)). Note that, as described above, v_(LX) andu_(LX) are set so as to satisfy the ratio of the average power 1:w².

Although the case of three code lengths is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u_(LX) for power change existwhen there are two or more code lengths that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(LX) for power change when the codelength is set, and performs power change. Another important point isthat two or more values v_(LX) for power change exist when there are twoor more code lengths that can be set, and the transmission deviceselects any of the values for power change from among the two or morevalues v_(LX) for power change when the code length is set, and performspower change.

Example 6-2

The following describes a scheme of setting the average power of s1 ands2 according to a coding rate for the error correction codes used togenerate s1 and s2 when the transmission device supports a plurality ofcoding rates for the error correction codes.

Examples of the error correction codes include block codes such as turbocodes or duo-binary turbo codes using tail-biting, LDPC codes, or thelike. In many communication systems and broadcasting systems, aplurality of coding rates are supported. Encoded data for which errorcorrection codes whose coding rate is selected from among the pluralityof supported coding rates has been performed is distributed to twosystems. The encoded data having been distributed to the two systems ismodulated in the modulation scheme for s1 and in the modulation schemefor s2 to generate the (mapped) baseband signals s1(t) and s2(t).

The control signal (8400) is a signal indicating the selected codingrate for the error correction codes described above. The power changer(8401A) sets the value v for power change according to the controlsignal (8400). Similarly, the power changer (8401B) sets the value u forpower change according to the control signal (8400).

The present invention is characterized in that the power changers (8401Aand 8401B) respectively set the values v and u for power changeaccording to the selected coding rate indicated by the control signal(8400). Here, values for power change set according to the coding raterx are referred to as v_(rx) and u_(rx).

For example, when r1 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r1). When r2 is selected asthe coding rate, the power changer (8401A) sets a value for power changeto v_(r2). When r3 is selected as the coding rate, the power changer(8401A) sets a value for power change to v_(r3).

Also, when r1 is selected as the coding rate, the power changer (8401B)sets a value for power change to u_(r1). When r2 is selected as thecoding rate, the power changer (8401B) sets a value for power change tou_(r3). When r3 is selected as the coding rate, the power changer(8401B) sets a value for power change to u_(r3).

In this case, for example, by setting v_(r1), v_(r2) and v_(r3) so as tobe different from one another, a high error correction capability can beachieved for each code length. Similarly, by setting u_(r1), u_(r2) andu_(r3) so as to be different from one another, a high error correctioncapability can be achieved for each coding rate. Depending on the setcoding rate, however, the effect might not be obtained even if the valuefor power change is changed. In such a case, even when the coding rateis changed, it is unnecessary to change the value for power change (forexample, v_(r1)=V_(r2) may be satisfied, and u_(r1)=u_(r3) may besatisfied. What is important is that two or more values exist in a setof v_(r1), v_(r2) and v_(r3), and that two or more values exist in a setof urn, u_(r3) and u_(r3)). Note that, as described above, v_(rX) andu_(rX) are set so as to satisfy the ratio of the average power 1:w².

Also, note that, as examples of r1, r2 and r3 described above, codingrates 1/2, 2/3 and 3/4 are considered when the error correction code isthe LDPC code.

Although the case of three coding rates is taken as an example in theabove description, the present invention is not limited to this. Oneimportant point is that two or more values u, for power change existwhen there are two or more coding rates that can be set, and thetransmission device selects any of the values for power change fromamong the two or more values u_(rx) for power change when the codingrate is set, and performs power change. Another important point is thattwo or more values v_(rX) for power change exist when there are two ormore coding rates that can be set, and the transmission device selectsany of the values for power change from among the two or more valuesv_(rX) for power change when the coding rate is set, and performs powerchange.

Example 6-3

In order for the reception device to achieve excellent data receptionquality, it is important to implement the following.

The following describes a scheme of setting the average power (averagevalues) of s1 and s2 according to a modulation scheme used to generates1 and s2 when the transmission device supports a plurality ofmodulation schemes.

Here, as an example, a case where the modulation scheme for s1 is fixedto 16QAM and the modulation scheme for s2 is changed from 64QAM to QPSKby the control signal (or can be set to either 16QAM or QPSK) isconsidered. In a case where the modulation scheme for s1 is 16QAM, themapping scheme for s1(t) is as shown in FIG. 80, and g is represented byformula 79 in FIG. 80. In a case where the modulation scheme for s2 is64QAM, the mapping scheme for s2(t) is as shown in FIG. 86, and k isrepresented by formula 85 in FIG. 86. Also, in a case where themodulation scheme for s2(t) is QPSK, the mapping scheme for s2(t) is asshown in FIG. 81, and h is represented by formula 78 in FIG. 81.

In FIG. 85, when the modulation scheme for s1 is 16QAM and themodulation scheme for s2 is 64QAM, assume that v=α and u=α×w₆₄. In thiscase, the ratio between the average power of 64QAM and the average powerof 16QAM is v²:u²=α²:α²×w₆₄ ²=1:w₆₄ ².

In FIG. 85, when the modulation scheme for s1 is 16QAM and themodulation scheme for s2 is QPSK, assume that v=β and u=β×w₄. In thiscase, the ratio between the average power of 64QAM and the average powerof QPSK is v²:u²=β²:β²×w₄ ²=1:w₄ ². In this case, according to theminimum Euclidean distance relationship, the reception device achieves ahigh data reception quality when w₄<w₆₄, regardless of whether themodulation scheme for s2 is 64QAM or QPSK.

Note that although “the modulation scheme for s1 is fixed to 16QAM” inthe description above, it is possible that “the modulation scheme for s2is fixed to 16QAM and the modulation scheme for s1 is changed from 64QAMto QPSK (set to either 16QAM or QPSK)”, w₄<w₆₄ should be fulfilled. (Thesame as described in Example 4-3). (Note that the value used for themultiplication for the power change in the case of 16QAM is u=α×w₁₆, thevalue used for the multiplication for the power change in the case ofQPSK is u=β×w₄, the value used for the power change in the case of 64QAMis v=α when the selectable modulation scheme is 16QAM and v=β when theselectable modulation scheme is QPSK). Also, when the set of (themodulation scheme for s1, the modulation scheme for s2) is selectablefrom the sets of (16QAM, 64QAM), (64QAM, 16QAM), (16QAM, QPSK) and(QPSK, 16QAM), w₄<w₆₄ should be fulfilled.

The following describes a case where the above-mentioned description isgeneralized.

For generalization, assume that the modulation scheme for s1 is fixed toa modulation scheme C with which the number of signal points in the IQplane is c. Also assume that the modulation scheme for s2 is selectablefrom a modulation scheme A with which the number of signal points in theIQ plane is a and a modulation scheme B with which the number of signalpoints in the IQ plane is b (c>b>a). In this case, when the modulationscheme for s2 is set to the modulation scheme A, assume that ratiobetween the average power of the modulation scheme for s1, which is themodulation scheme C, and the average power of the modulation scheme fors2, which is the modulation scheme A, is 1:w_(a) ². Also, when themodulation scheme for s2 is set to the modulation scheme B, assume thatratio between the average power of the modulation scheme for s1, whichis the modulation scheme C, and the average power of the modulationscheme for s2, which is the modulation scheme B, is 1:w_(b) ². If thisis the case, the reception device achieves a high data reception qualitywhen w_(a)<w_(b) is fulfilled.

Note that although “the modulation scheme for s1 is fixed to C” in thedescription above, even when “the modulation scheme for s2 is fixed tothe modulation scheme C and the modulation scheme for s1 is changed fromthe modulation scheme A to the modulation scheme B (set to either themodulation scheme A or the modulation scheme B), the average powersshould fulfill w_(a)<w_(b). (If this is the case, as with thedescription above, when the average power of the modulation scheme is C,the average power of the modulation scheme A is w_(a) ², and the averagepower of the modulation scheme B is w_(b) ².) Also, when the set of (themodulation scheme for s1 and the modulation scheme for s2) is selectablefrom the sets of (the modulation scheme C and the modulation scheme A),(the modulation scheme A and the modulation scheme C), (the modulationscheme C and the modulation scheme B) and (the modulation scheme B andthe modulation scheme C), the average powers should fulfill w_(a)<w_(b).

In the present description including “Embodiment 1”, and so on, thepower consumption by the transmission device can be reduced by settingα=1 in the formula 36 representing the precoding matrices used for thescheme for regularly changing the phase. This is because the averagepower of z1 and the average power of z2 are the same even when “theaverage power (average value) of s1 and the average power (averagevalue) of s2 are set to be different when the modulation scheme for s1and the modulation scheme for s2 are different”, and setting α=1 doesnot result in increasing the PAPR (Peak-to-Average Power Ratio) of thetransmission power amplifier provided in the transmission device.

However, even when α≠1, there are some precoding matrices that can beused with the scheme that regularly changes the phase and have limitedinfluence to PAPR. For example, when the precoding matrices representedby formula 36 in Embodiment 1 are used to achieve the scheme forregularly changing the phase, the precoding matrices have limitedinfluence to PAPR even when α≠1.

Operations of the Reception Device

Subsequently, explanation is provided of the operations of the receptiondevice. Explanation of the reception device has already been provided inEmbodiment 1 and so on, and the structure of the reception device isillustrated in FIGS. 7, 8 and 9, for instance

According to the relation illustrated in FIG. 5, when the transmissiondevice transmits modulated signals as introduced in FIGS. 84 and 85, onerelation among the two relations denoted by the two formulas below issatisfied. Note that in the two formulas below, r1(t) and r2(t) indicatereception signals, and h11(t), h12(t), h21(t), and h22(t) indicatechannel fluctuation values.

In the case of Example 1, Example 2 and Example 3, the followingrelationship shown in formula 89 is derived from FIG. 5.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 89} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{r\; 2(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}e^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}1 & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{s\; 1(t)} \\{{us}\; 2(t)}\end{pmatrix}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 89} \right)\end{matrix}$

Also, as explained in Example 1, Example 2, and Example 3, therelationship may be as shown in formula 90 below:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 90} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{r\; 2(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ue}^{j\; 0} & 0 \\0 & e^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}u & 0 \\0 & 1\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{{us}\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 90} \right)\end{matrix}$

The reception device performs demodulation (detection) (i.e. estimatesthe bits transmitted by the transmission device) by using therelationships described above (in the same manner as described inEmbodiment 1 and so on).

In the case of Example 4, Example 5 and Example 6, the followingrelationship shown in formula 91 is derived from FIG. 5.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 91} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{{r2}(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & {v \times w}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{{vs}\; 1(t)} \\{{us}\; 2(t)}\end{pmatrix}}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{{vs}\; 1(t)} \\{v \times w \times s\; 2(t)}\end{pmatrix}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 91} \right)\end{matrix}$

Also, as explained in Example 3, Example 4, and Example 5, therelationship may be as shown in formula 92 below:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 92} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{{r2}(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ue}^{j\; 0} & 0 \\0 & {ve}^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{v \times w} & 0 \\0 & v\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{{us}\; 1(t)} \\{{vs}\; 2(t)}\end{pmatrix}}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{v \times {ws}\; 1(t)} \\{{vs}\; 2(t)}\end{pmatrix}}}}\end{matrix} & \left( {{formula}\mspace{14mu} 92} \right)\end{matrix}$

The reception device performs demodulation (detection) (i.e. estimatesthe bits transmitted by the transmission device) by using therelationships described above (in the same manner as described inEmbodiment 1 and so on).

Note that although Examples 1 through 6 show the case where the powerchanger is added to the transmission device, the power change may beperformed at the stage of mapping.

As described in Example 1, Example 2, and Example 3, and as particularlyshown in formula 89, the mapper 306B in FIG. 3 and FIG. 4 may outputu×s2(t), and the power changer may be omitted in such cases. If this isthe case, it can be said that the scheme for regularly changing thephase is applied to the signal s1(t) after the mapping and the signalu×s2(t) after the mapping, the modulated signal after precoding.

As described in Example 1, Example 2, and Example 3, and as particularlyshown in formula 90, the mapper 306A in FIG. 3 and FIG. 4 may outputu×s1(t), and the power changer may be omitted in such cases. If this isthe case, it can be said that the scheme for regularly changing thephase is applied to the signal s2(t) after the mapping and the signalu×s1(t) after the mapping, the modulated signal after precoding.

In Example 4, Example 5, and Example 6, as particularly shown in formula91, the mapper 306A in FIG. 3 and FIG. 4 may output v=s1(t), and themapper 306B may output u×s2(t), and the power changer may be omitted insuch cases. If this is the case, it can be said that the scheme forregularly changing the phase is applied to the signal v=s1(t) after themapping and the signal u×s2(t) after the mapping, the modulated signalsafter precoding.

In Example 4, Example 5, and Example 6, as particularly shown in formula92, the mapper 306A in FIG. 3 and FIG. 4 may output u×s1(t), and themapper 306B may output v×s2(t), and the power changer may be omitted insuch cases. If this is the case, it can be said that the scheme forregularly changing the phase is applied to the signal u×s1(t) after themapping and the signal v×s2(t) after the mapping, the modulated signalsafter precoding.

Note that F shown in formulas 89 through 92 denotes precoding matricesused at time t, and y(t) denotes phase changing values. The receptiondevice performs demodulation (detection) by using the relationshipsbetween r1(t), r2(t) and s1(t), s2(t) described above (in the samemanner as described in Embodiment 1 and so on). However, distortioncomponents, such as noise components, frequency offset, channelestimation error, and the likes are not considered in the formulasdescribed above. Hence, demodulation (detection) is performed with them.Regarding the values u and v that the transmission device uses forperforming the power change, the transmission device transmitsinformation about these values, or transmits information of thetransmission mode (such as the transmission scheme, the modulationscheme and the error correction scheme) to be used. The reception devicedetects the values used by the transmission device by acquiring theinformation, obtains the relationships described above, and performs thedemodulation (detection).

In the present embodiment, the switching between the phase changingvalues is performed on the modulated signal after precoding in the timedomain. However, when a multi-carrier transmission scheme such as anOFDM scheme is used, the present invention is applicable to the casewhere the switching between the phase changing values is performed onthe modulated signal after precoding in the frequency domain, asdescribed in other embodiments. If this is the case, t used in thepresent embodiment is to be replaced with f (frequency ((sub) carrier)).

Accordingly, in the case of performing the switching between the phasechanging values on the modulated signal after precoding in the timedomain, z1(t) and z2(t) at the same time point is transmitted fromdifferent antennas by using the same frequency. On the other hand, inthe case of performing the switching between the phase changing valueson the modulated signal after precoding in the frequency domain, z1(f)and z2(f) at the same frequency is transmitted from different antennasat the same time point.

Also, even in the case of performing switching between the phasechanging values on the modulated signal after precoding in the time andfrequency domains, the present invention is applicable as described inother embodiments. The scheme pertaining to the present embodiment,which switches between the phase changing values on the modulated signalafter precoding, is not limited the scheme which switches between thephase changing values on the modulated signal after precoding asdescribed in the present Description.

Also, assume that processed baseband signals z1(i), z2(i) (where irepresents the order in terms of time or frequency (carrier)) aregenerated by regular phase change and precoding (it does not matterwhich is performed first) on baseband signals s1(i) and s2(i) for twostreams. Let the in-phase component I and the quadrature component Q ofthe processed baseband signal z1(i) be I₁(i) and Q₁(i) respectively, andlet the in-phase component I and the quadrature component Q of theprocessed baseband signal z2(i) be I₂(i) and Q₂(i) respectively. In thiscase, the baseband components may be switched, and modulated signalscorresponding to the switched baseband signal r1(i) and the switchedbaseband signal r2(i) may be transmitted from different antennas at thesame time and over the same frequency by transmitting a modulated signalcorresponding to the switched baseband signal r1(i) from transmitantenna 1 and a modulated signal corresponding to the switched basebandsignal r2(i) from transmit antenna 2 at the same time and over the samefrequency. Baseband components may be switched as follows.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i) and Q₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be I₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i) and I₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₁(i) and Q₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₁(i) and Q₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i) and I₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i) and Q₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₁(i) and I₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be Q₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be I₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be Q₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be Q₁(i) and I₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i) and I₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be Q₁(i) and Q₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr₁(i) be Q₁(i) and Q₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i) and I₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be Q₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be Q₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i) and Q₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be I₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i) and Q₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be Q₁(i) and I₂(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be Q₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be I₂(i) and Q₁(i) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be Q₂(i) and I₁(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr1(i) be Q₁(i) and I₂(i) respectively.

In the above description, signals in two streams are processed andin-phase components and quadrature components of the processed signalsare switched, but the present invention is not limited in this way.Signals in more than two streams may be processed, and the in-phasecomponents and quadrature components of the processed signals may beswitched.

In addition, the signals may be switched in the following manner. Forexample,

Let the in-phase component and the quadrature component of the switchedbaseband signal r1 (i) be I₂(i) and Q₂(i) respectively, and the in-phasecomponent and the quadrature component of the switched baseband signalr2(i) be I₁(i) and Q₁(i) respectively.

Such switching can be achieved by the structure shown in FIG. 55.

In the above-mentioned example, switching between baseband signals atthe same time (at the same frequency ((sub)carrier)) has been described,but the present invention is not limited to the switching betweenbaseband signals at the same time. As an example, the followingdescription can be made.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i+v) and Q₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be I₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i+v) and I₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₁(i+v) and Q₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₁(i+v) and Q₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i+v) and I₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₁(i+v) and Q₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₁(i+v) and I₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be Q₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be I₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be Q₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be Q₁(i+v) and I₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i+v) and I₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₁(i+v) and Q₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₁(i+v) and Q₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i+v) and I₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i+v) and Q₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be I₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be I₁(i+v) and Q₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₁(i+v) and I₂(i+w) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be Q₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be I₂(i+w) and Q₁(i+v) respectively.

Let the in-phase component and the quadrature component of the switchedbaseband signal r2(i) be Q₂(i+w) and I₁(i+v) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r1(i) be Q₁(i+v) and I₂(i+w) respectively.

In addition, the signals may be switched in the following manner. Forexample,

Let the in-phase component and the quadrature component of the switchedbaseband signal r1(i) be I₂(i+w) and Q₂(i+w) respectively, and thein-phase component and the quadrature component of the switched basebandsignal r2(i) be I₁(i+v) and Q₁(i+w) respectively.

This can also be achieved by the structure shown in FIG. 55.

FIG. 55 illustrates a baseband signal switcher 5502 explaining theabove. As shown, of the two processed baseband signals z1(i) 5501_1 andz2(i) 5501_2, processed baseband signal z1(i) 5501_1 has in-phasecomponent I₁(i) and quadrature component Q₁(i), while processed basebandsignal z2(i) 55012 has in-phase component I₂(i) and quadrature componentQ₂(i). Then, after switching, switched baseband signal r1(i) 5503_1 hasin-phase component I_(r1)(i) and quadrature component Q_(r1)(i), whileswitched baseband signal r2(i) 5503_2 has in-phase component I₂(i) andquadrature component Q₂(i). The in-phase component I_(r1)(i) andquadrature component Q_(r1)(i) of switched baseband signal r1(i) 5503_1and the in-phase component I_(r2)(i) and quadrature component Q_(r2)(i)of switched baseband signal r2(i) 5503_2 may be expressed as any of theabove. Although this example describes switching performed on basebandsignals having a common time (common ((sub-)carrier) frequency) andhaving undergone two types of signal processing, the same may be appliedto baseband signals having undergone two types of signal processing buthaving different time (different ((sub-)carrier) frequencies).

The switching may be performed while regularly changing switchingmethods.

For example,

At time 0,

for switched baseband signal r1(0), the in-phase component may be I₁(0)while the quadrature component may be Q₁(0), and for switched basebandsignal r2(0), the in-phase component may be I₂(0) while the quadraturecomponent may be Q₂(0);

At time 1,

for switched baseband signal r1(1), the in-phase component may be I₂(1)while the quadrature component may be Q₂(1), and for switched basebandsignal r2(1), the in-phase component may be I₁(1) while the quadraturecomponent may be Q₁(1), and so on. In other words,

When time is 2k (k is an integer),

for switched baseband signal r1(2k), the in-phase component may beI₁(2k) while the quadrature component may be Q₁(2k), and for switchedbaseband signal r2(2k), the in-phase component may be I₂(2k) while thequadrature component may be Q₂(2k).

When time is 2k+1 (k is an integer),

for switched baseband signal r1(2k+1), the in-phase component may beI₂(2k+1) while the quadrature component may be Q₂(2k+1), and forswitched baseband signal r2(2k+1), the in-phase component may beI₁(2k+1) while the quadrature component may be Q₁(2k+1).

When time is 2k (k is an integer),

for switched baseband signal r1(2k), the in-phase component may beI₂(2k) while the quadrature component may be Q₂(2k), and for switchedbaseband signal r2(2k), the in-phase component may be I₁(2k) while thequadrature component may be Q₁(2k).

When time is 2k+1 (k is an integer),

for switched baseband signal r1(2k+1), the in-phase component may beI₁(2k+1) while the quadrature component may be Q₁(2k+1), and forswitched baseband signal r2(2k+1), the in-phase component may beI₂(2k+1) while the quadrature component may be Q₂(2k+1).

Similarly, the switching may be performed in the frequency domain. Inother words,

When frequency ((sub) carrier) is 2k (k is an integer),

for switched baseband signal r1(2k), the in-phase component may beI₁(2k) while the quadrature component may be Q₁(2k), and for switchedbaseband signal r2(2k), the in-phase component may be I₂(2k) while thequadrature component may be Q₂(2k).

When frequency ((sub) carrier) is 2k+1 (k is an integer),

for switched baseband signal r1(2k+1), the in-phase component may beI₂(2k+1) while the quadrature component may be Q₂(2k+1), and forswitched baseband signal r2(2k+1), the in-phase component may beI₁(2k+1) while the quadrature component may be Q₁(2k+1).

When frequency ((sub) carrier) is 2k (k is an integer),

for switched baseband signal r1(2k), the in-phase component may beI₂(2k) while the quadrature component may be Q₂(2k), and for switchedbaseband signal r2(2k), the in-phase component may be I₁(2k) while thequadrature component may be Q₁(2k).

When frequency ((sub) carrier) is 2k+1 (k is an integer),

for switched baseband signal r1(2k+1), the in-phase component may beI₁(2k+1) while the quadrature component may be Q₁(2k+1), and forswitched baseband signal r2(2k+1), the in-phase component may beI₂(2k+1) while the quadrature component may be Q₂(2k+1).

Embodiment G1

The present embodiment describes a scheme that is used when themodulated signal subject to the QPSK mapping and the modulated signalsubject to the 16QAM mapping are transmitted, for example, and is usedfor setting the average power of the modulated signal subject to theQPSK mapping and the average power of the modulated signal subject tothe 16QAM mapping such that the average powers will be different fromeach other. This scheme is different from Embodiment F1.

As explained in Embodiment F1, when the modulation scheme for themodulated signal of s1 is QPSK and the modulation scheme for themodulated signal of s2 is 16QAM (or the modulation scheme for themodulated signal s1 is 16QAM and the modulation scheme for the modulatedsignal s2 is QPSK), if the average power of the modulated signal subjectto the QPSK mapping and the average power of the modulated signalsubject to the 16QAM mapping are set to be different from each other,the PAPR (Peak-to-Average Power Ratio) of the transmission poweramplifier provided in the transmission device may increase, depending onthe precoding matrix used by the transmission device. The increase ofthe PAPR may lead to the increase in power consumption by thetransmission device.

In the present embodiment, description is provided on the scheme forregularly performing phase change after performing the precodingdescribed in “Embodiment 1” and so on, where, even when α≠1 in theformula 36 of the precoding matrix to be used in the scheme forregularly changing the phase, the influence to the PAPR is suppressed toa minimal extent.

In the present embodiment, description is provided taking as an examplea case where the modulation scheme applied to the streams s1 and s2 iseither QPSK or 16QAM.

Firstly, explanation is provided of the mapping scheme for QPSKmodulation and the mapping scheme for 16QAM modulation. Note that, inthe present embodiment, the symbols s1 and s2 refer to signals which areeither in accordance with the mapping for QPSK modulation or the mappingfor 16QAM modulation.

First of all, description is provided concerning mapping for 16QAM withreference to the accompanying FIG. 80. FIG. 80 illustrates an example ofa signal point layout in the I-Q plane (I: in-phase component; Q:quadrature component) for 16QAM. Concerning the signal point 9400 inFIG. 94, when the bits transferred (input bits) are b0-b3, that is, whenthe bits transferred are indicated by (b0, b1, b2, b3)=(1, 0, 0, 0)(this value being illustrated in FIG. 94), the coordinates in the I-Qplane (I: in-phase component; Q: quadrature component) correspondingthereto is denoted as (I, Q)=(−3×g, 3×g). The values of coordinates Iand Q in this set of coordinates indicates the mapped signals. Notethat, when the bits transferred (b0, b1, b2, b3) take other values thanin the above, the set of values I and Q is determined according to thevalues of the bits transferred (b0, b1, b2, b3) and according to FIG.80. Further, similar as in the above, the values of coordinates I and Qin this set indicates the mapped signals (s1 and s2).

Subsequently, description is provided concerning mapping for QPSKmodulation with reference to the accompanying FIG. 81. FIG. 81illustrates an example of a signal point layout in the I-Q plane (I:in-phase component; Q: quadrature component) for QPSK. Concerning thesignal point 8100 in FIG. 81, when the bits transferred (input bits) areb0 and b1, that is, when the bits transferred are indicated by (b0,b1)=(1, 0) (this value being illustrated in FIG. 81), the coordinates inthe I-Q plane (I: in-phase component; Q: quadrature component)corresponding thereto is denoted as (I, Q)=(−1×h, 1×h). Further, thevalues of coordinates I and Q in this set of coordinates indicates themapped signals. Note that, when the bits transferred (b0, b1) take othervalues than in the above, the set of coordinates (I, Q) is determinedaccording to the values of the bits transferred (b0, b1) and accordingto FIG. 81. Further, similar as in the above, the values of coordinatesI and Q in this set indicates the mapped signals (s1 and s2).

Further, when the modulation scheme applied to s1 and s2 is either QPSKor 16QAM, in order to equalize the values of the average power, h is asrepresented by formula 78, and g is as represented by formula 79.

FIGS. 87 and 88 illustrate an example of the scheme of changing themodulation scheme, the power changing value, and the precoding matrix inthe time domain (or in the frequency domain, or in the time domain andthe frequency domain) when using a precoding-related signal processorillustrated in FIG. 85.

In FIG. 87, a chart is provided indicating the modulation scheme, thepower changing value (u, v), and the phase changing value (y[t]) to beset at each of times t=0 through t=11. Note that, concerning themodulated signals z1(t) and z2(t), the modulated signals z1(t) and z2(t)at the same time point are to be simultaneously transmitted fromdifferent transmit antennas at the same frequency. (Although the chartin FIG. 87 is based on the time domain, when using a multi-carriertransmission scheme as the OFDM scheme, switching between schemes(modulation scheme, power changing value, phase changing value) may beperformed according to the frequency (subcarrier) domain, rather thanaccording to the time domain. In such a case, replacement should be madeof t=0 with f=f0, t=1 with f=f1, . . . , as is shown in FIG. 87. (Notethat here, f denotes frequencies (subcarriers), and thus, f0, f1, . . ., indicate different frequencies (subcarriers) to be used.) Further,note that concerning the modulated signals z1(f) and z2(f) in such acase, the modulated signals z1(f) and z2(f) having the same frequencyare to be simultaneously transmitted from different transmit antennas.

As illustrated in FIG. 87, when the modulation scheme applied is QPSK,the power changer (although referred to as the power changer herein, mayalso be referred to as an amplification changer or a weight unit)multiplies a (a being a real number) with respect to a signal modulatedin accordance with QPSK. Similarly, when the modulation scheme appliedis 16QAM, the power changer (although referred to as the power changerherein, may also be referred to as the amplification changer or theweight unit) multiplies b (b being a real number) with respect to asignal modulated in accordance with 16QAM.

In the example illustrated in FIG. 87, three phase changing values,namely y[0], y[1], and y[2] are prepared as phase changing values usedin the scheme for regularly performing phase change after precoding.Additionally, the period (cycle) for the scheme for regularly performingphase change after precoding is 3 (thus, each of t0-t2, t3-t5, . . .composes one period (cycle)). Note, in this embodiment, since the phasechange is performed on one of the signals after precoding as shown inthe example in FIG. 85, y[i] is an imaginary number having the absolutevalue of 1 (i.e. y[i]=e^(jθ)). However, as described in thisDescription, the phase change may be performed after performing theprecoding on a plurality of signals. If this is the case, a phasechanging value exists for each of the plurality of signals afterprecoding.

The modulation scheme applied to s1(t) is QPSK in period (cycle) t0-t2,16QAM in period (cycle) t3-t5 and so on, whereas the modulation schemeapplied to s2(t) is 16QAM in period (cycle) t0-t2, QPSK in period(cycle) t3-t5 and so on. Thus, the set of (modulation scheme of s1(t),modulation scheme of s2(t)) is either (QPSK, 16QAM) or (16QAM, QPSK).

Here, it is important that:

when performing phase change according to y[0], both (QPSK, 16QAM) and(16QAM, QPSK) can be the set of (modulation scheme of s1(t), modulationscheme of s2(t)), when performing phase change according to y[1], both(QPSK, 16QAM) and (16QAM, QPSK) can be the set of (modulation scheme ofs1(t), modulation scheme of s2(t)), and similarly, when performing phasechange according to y[2], both (QPSK, 16QAM) and (16QAM, QPSK) can bethe set of (modulation scheme of s1(t), modulation scheme of s2(t)).

In addition, when the modulation scheme applied to s1(t) is QPSK, thepower changer (8501A) multiples s1(t) with a and thereby outputsa×s1(t). On the other hand, when the modulation scheme applied to s1(t)is 16QAM, the power changer (8501A) multiples s1(t) with b and therebyoutputs b× s1(t).

Further, when the modulation scheme applied to s2(t) is QPSK, the powerchanger (8501B) multiples s2(t) with a and thereby outputs a×s2(t). Onthe other hand, when the modulation scheme applied to s2(t) is 16QAM,the power changer (8501B) multiples s2(t) with b and thereby outputs b×s2(t).

Note that, regarding the scheme for differently setting the averagepower of signals in accordance with mapping for QPSK modulation and theaverage power of signals in accordance with mapping for 16QAMmodulation, description has already been made in Embodiment F1.

Thus, when taking the set of (modulation scheme of s1(t), modulationscheme of s2(t)) into consideration, the period (cycle) for the phasechange and the switching between modulation schemes is 6=3×2 (where 3:the number of phase changing values prepared as phase changing valuesused in the scheme for regularly performing phase change afterprecoding, and 2: both (QPSK, 16QAM) and (16QAM, QPSK) can be the set of(modulation scheme of s1(t), modulation scheme of s2(t)) for each of thephase changing values), as shown in FIG. 87.

As description has been made in the above, by making an arrangement suchthat both (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of(modulation scheme of s1(t), modulation scheme of s2(t)), and such thatboth (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of (modulationscheme of s1(t), modulation scheme of s2(t)) with respect to each of thephase changing values prepared as phase changing values used in thescheme for regularly performing phase change, the following advantageouseffects are to be yielded. That is, even when differently setting theaverage power of signals in accordance with mapping for QPSK modulationand the average power of signals in accordance with mapping for 16QAMmodulation, the influence with respect to the PAPR of the transmissionpower amplifier included in the transmission device is suppressed to aminimal extent, and thus the influence with respect to the powerconsumption of the transmission device is suppressed to a minimalextent, while the reception quality of data received by the receptiondevice in the LOS environment is improved, as explanation has alreadybeen provided in the present description.

Note that, although description has been provided in the above, takingas an example a case where the set of (modulation scheme of s1(t),modulation scheme of s2(t)) is (QPSK, 16QAM) and (16QAM, QPSK), thepossible sets of (modulation scheme of s1(t), modulation scheme ofs2(t)) are not limited to this. More specifically, the set of(modulation scheme of s1(t), modulation scheme of s2(t)) may be one of:(QPSK, 64QAM), (64QAM, QPSK); (16QAM, 64QAM), (64QAM, 16QAM); (128QAM,64QAM), (64QAM, 128QAM); (256QAM, 64QAM), (64QAM, 256QAM), and the like.That is, the present invention is to be similarly implemented providedthat two different modulation schemes are prepared, and a different oneof the modulation schemes is applied to each of s1(t) and s2(t).

In FIG. 88, a chart is provided indicating the modulation scheme, thepower changing value, and the phase changing value to be set at each oftimes t=0 through t=11. Note that, concerning the modulated signalsz1(t) and z2(t), the modulated signals z1(t) and z2(t) at the same timepoint are to be simultaneously transmitted from different transmitantennas at the same frequency. (Although the chart in FIG. 88 is basedon the time domain, when using a multi-carrier transmission scheme asthe OFDM scheme, switching between schemes may be performed according tothe frequency (subcarrier) domain, rather than according to the timedomain. In such a case, replacement should be made of t=0 with f=f0, t=1with f=f1, . . . , as is shown in FIG. 88. (Note that here, f denotesfrequencies (subcarriers), and thus, f0, f1, . . . , indicate differentfrequencies (subcarriers) to be used.) Further, note that concerning themodulated signals z1(f) and z2(f) in such a case, the modulated signalsz1(f) and z2(f) having the same frequency are to be simultaneouslytransmitted from different transmit antennas. Note that the exampleillustrated in FIG. 88 is an example that differs from the exampleillustrated in FIG. 87, but satisfies the requirements explained withreference to FIG. 87.

As illustrated in FIG. 88, when the modulation scheme applied is QPSK,the power changer (although referred to as the power changer herein, mayalso be referred to as an amplification changer or a weight unit)multiplies a (a being a real number) with respect to a signal modulatedin accordance with QPSK. Similarly, when the modulation scheme appliedis 16QAM, the power changer (although referred to as the power changerherein, may also be referred to as the amplification changer or theweight unit) multiplies b (b being a real number) with respect to asignal modulated in accordance with 16QAM.

In the example illustrated in FIG. 88, three phase changing values,namely y[0], y[1], and y[2] are prepared as phase changing values usedin the scheme for regularly performing phase change after precoding.Additionally, the period (cycle) for the scheme for regularly performingphase change after precoding is 3 (thus, each of t0-t2, t3-t5, . . .composes one period (cycle)).

Further, QPSK and 16QAM are alternately set as the modulation schemeapplied to s1(t) in the time domain, and the same applies to themodulation scheme set to s2(t). The set of (modulation scheme of s1(t),modulation scheme of s2(t)) is either (QPSK, 16QAM) or (16QAM, QPSK).

Here, it is important that: when performing phase change according toy[0], both (QPSK, 16QAM) and (16QAM, QPSK) can be the set of (modulationscheme of s1(t), modulation scheme of s2(t)), when performing phasechange according to y[1], both (QPSK, 16QAM) and (16QAM, QPSK) can bethe set of (modulation scheme of s1(t), modulation scheme of s2(t)), andsimilarly, when performing phase change according to y[2], both (QPSK,16QAM) and (16QAM, QPSK) can be the set of (modulation scheme of s1(t),modulation scheme of s2(t)).

In addition, when the modulation scheme applied to s1(t) is QPSK, thepower changer (8501A) multiples s1(t) with a and thereby outputsa×s1(t). On the other hand, when the modulation scheme applied to s1(t)is 16QAM, the power changer (8501A) multiples s1(t) with b and therebyoutputs b× s1(t).

Further, when the modulation scheme applied to s2(t) is QPSK, the powerchanger (8501B) multiples s2(t) with a and thereby outputs a×s2(t). Onthe other hand, when the modulation scheme applied to s2(t) is 16QAM,the power changer (8501B) multiples s2(t) with b and thereby outputsb×s2(t).

Thus, when taking the set of (modulation scheme of s1(t), modulationscheme of s2(t)) into consideration, the period (cycle) for the phasechange and the switching between modulation schemes is 6=3×2 (where 3:the number of phase changing values prepared as phase changing valuesused in the scheme for regularly performing phase change afterprecoding, and 2: both (QPSK, 16QAM) and (16QAM, QPSK) can be the set of(modulation scheme of s1(t), modulation scheme of s2(t)) for each of thephase changing values), as shown in FIG. 88.

As description has been made in the above, by making an arrangement suchthat both (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of(modulation scheme of s1(t), modulation scheme of s2(t)), and such thatboth (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of (modulationscheme of s1(t), modulation scheme of s2(t)) with respect to each of thephase changing values prepared as phase changing values used in thescheme for regularly performing phase change, the following advantageouseffects are to be yielded. That is, even when differently setting theaverage power of signals in accordance with mapping for QPSK modulationand the average power of signals in accordance with mapping for 16QAMmodulation, the influence with respect to the PAPR of the transmissionpower amplifier included in the transmission device is suppressed to aminimal extent, and thus the influence with respect to the powerconsumption of the transmission device is suppressed to a minimalextent, while the reception quality of data received by the receptiondevice in the LOS environment is improved, as explanation has alreadybeen provided in the present description.

Note that, although description has been provided in the above, takingas an example a case where the set of (modulation scheme of s1(t),modulation scheme of s2(t)) is (QPSK, 16QAM) and (16QAM, QPSK), thepossible sets of (modulation scheme of s1(t), modulation scheme ofs2(t)) are not limited to this. More specifically, the set of(modulation scheme of s1(t), modulation scheme of s2(t)) may be one of:(QPSK, 64QAM), (64QAM, QPSK); (16QAM, 64QAM), (64QAM, 16QAM); (128QAM,64QAM), (64QAM, 128QAM); (256QAM, 64QAM), (64QAM, 256QAM), and the like.That is, the present invention is to be similarly implemented providedthat two different modulation schemes are prepared, and a different oneof the modulation schemes is applied to each of s1(t) and s2(t).

Additionally, the relation between the modulation scheme, the powerchanging value, and the phase changing value set at each of times (orfor each of frequencies) is not limited to those described in the abovewith reference to FIGS. 87 and 88.

To summarize the explanation provided in the above, the following pointsare essential.

Arrangements are to be made such that the set of (modulation scheme ofs1(t), modulation scheme of s2(t)) can be either (modulation scheme A,modulation scheme B) or (modulation scheme B, modulation scheme A), andsuch that the average power of signals in accordance with mapping forQPSK modulation and the average power of signals in accordance withmapping for 16QAM modulation are differently set. Further, when themodulation scheme applied to s1(t) is modulation scheme A, the powerchanger (8501A) multiples s1(t) with a and thereby outputs a×s1(t).Further, when the modulation scheme applied to s1(t) is modulationscheme B, the power changer (8501A) multiples s1(t) with a and therebyoutputs b×s1(t). Similarly, when the modulation scheme applied to s2(t)is modulation scheme A, the power changer (8501B) multiples s2(t) with aand thereby outputs a×s2(t). On the other hand, when the modulationscheme applied to s2(t) is modulation scheme B, the power changer(8501A) multiples s2(t) with b and thereby outputs b×s2(t).

Further, an arrangement is to be made such that phase changing valuesy[0], y[1], . . . , y[n−2], and y[n−1] (or y[k], where k satisfies0≤k≤n−1) exist as phase changing values prepared for use in the schemefor regularly performing phase change after precoding. Further, anarrangement is to be made such that both (modulation scheme A,modulation scheme B) and (modulation scheme B, modulation scheme A)exist as the set of (modulation scheme of s1(t), modulation scheme ofs2(t)) for y[k]. (Here, the arrangement may be made such that both(modulation scheme A, modulation scheme B) and (modulation scheme B,modulation scheme A) exist as the set of (modulation scheme of s1(t),modulation scheme of s2(t)) for y[k] for all values of k, or such that avalue k exists where both (modulation scheme A, modulation scheme B) and(modulation scheme B, modulation scheme A) exist as the set of(modulation scheme of s1(t), modulation scheme of s2(t)) for y[k].)

As description has been made in the above, by making an arrangement suchthat both (modulation scheme A, modulation scheme B) and (modulationscheme B, modulation scheme A) exist as the set of (modulation scheme ofs1(t), modulation scheme of s2(t)), and such that both (modulationscheme A, modulation scheme B) and (modulation scheme B, modulationscheme A) exist as the set of (modulation scheme of s1(t), modulationscheme of s2(t)) with respect to each of the phase changing valuesprepared as phase changing values used in the scheme for regularlyperforming phase change, the following advantageous effects are to beyielded. That is, even when differently setting the average power ofsignals for modulation scheme A and the average power of signals formodulation scheme B, the influence with respect to the PAPR of thetransmission power amplifier included in the transmission device issuppressed to a minimal extent, and thus the influence with respect tothe power consumption of the transmission device is suppressed to aminimal extent, while the reception quality of data received by thereception device in the LOS environment is improved, as explanation hasalready been provided in the present description.

In connection with the above, explanation is provided of a scheme forgenerating baseband signals s1(t) and s2(t) in the following. Asillustrated in FIGS. 3 and 4, the baseband signal s1(t) is generated bythe mapper 306A and the baseband signal s2(t) is generated by the mapper306B. As such, in the examples provided in the above with reference toFIGS. 87 and 88, the mapper 306A and 306B switch between mappingaccording to QPSK and mapping according to 16QAM by referring to thecharts illustrated in FIGS. 87 and 88.

Here, note that, although separate mappers for generating each of thebaseband signal s1(t) and the baseband signal s2(t) are provided in theillustrations in FIGS. 3 and 4, the present invention is not limited tothis. For instance, the mapper (8902) may receive input of digital data(8901), generate s1(t) and s2(t) according to FIGS. 87 and 88, andrespectively output s1(t) as the mapped signal 307A and s2(t) as themapped signal 307B.

FIG. 90 illustrates one structural example of the periphery of theweighting unit (precoding unit), which differs from the structuresillustrated in FIGS. 85 and 89. In FIG. 90, elements that operate in asimilar way to FIG. 3 and FIG. 85 bear the same reference signs. In FIG.91, a chart is provided indicating the modulation scheme, the powerchanging value, and the phase changing value to be set at each of timest=0 through t=11 with respect to the structural example illustrated inFIG. 90. Note that, concerning the modulated signals z1(t) and z2(t),the modulated signals z1(t) and z2(t) at the same time point are to besimultaneously transmitted from different transmit antennas at the samefrequency. (Although the chart in FIG. 91 is based on the time domain,when using a multi-carrier transmission scheme as the OFDM scheme,switching between schemes may be performed according to the frequency(subcarrier) domain, rather than according to the time domain. In such acase, replacement should be made of t=0 with f=f0, t=1 with f=f1, . . ., as is shown in FIG. 91. (Note that here, f denotes frequencies(subcarriers), and thus, f0, f1, . . . , indicate different frequencies(subcarriers) to be used.) Further, note that concerning the modulatedsignals z1(f) and z2(f) in such a case, the modulated signals z1(f) andz2(f) having the same frequency are to be simultaneously transmittedfrom different transmit antennas.

As illustrated in FIG. 91, when the modulation scheme applied is QPSK,the power changer (although referred to as the power changer herein, mayalso be referred to as an amplification changer or a weight unit)multiplies a (a being a real number) with respect to a signal modulatedin accordance with QPSK. Similarly, when the modulation scheme appliedis 16QAM, the power changer (although referred to as the power changerherein, may also be referred to as the amplification changer or theweight unit) multiplies b (b being a real number) with respect to asignal modulated in accordance with 16QAM.

In the example illustrated in FIG. 91, three phase changing values,namely y[0], y[1], and y[2] are prepared as phase changing values usedin the scheme for regularly performing phase change after precoding.Additionally, the period (cycle) for the scheme for regularly performingphase change after precoding is 3 (thus, each of t0-t2, t3-t5, . . .composes one period (cycle)).

Further, the modulation scheme applied to s1(t) is fixed to QPSK, andthe modulation scheme to be applied to s2(t) is fixed to 16QAM.Additionally, the signal switcher (9001) illustrated in FIG. 90 receivesthe mapped signals 307A and 307B and the control signal (8500) as inputthereto. The signal switcher (9001) performs switching with respect tothe mapped signals 307A and 307B according to the control signal (8500)(there are also cases where the switching is not performed), and outputsswitched signals (9002A: Ω1(t), and 9002B: Ω2(t)).

Here, it is important that:

When performing phase change according to y[0], both (QPSK, 16QAM) and(16QAM, QPSK) can be the set of (modulation scheme of Ω1(t), modulationscheme of Ω2(t)), when performing phase change according to y[1], both(QPSK, 16QAM) and (16QAM, QPSK) can be the set of (modulation scheme ofΩ1(t), modulation scheme of Ω2(t)), and similarly, when performing phasechange according to y[2], both (QPSK, 16QAM) and (16QAM, QPSK) can bethe set of (modulation scheme of Ω1(t), modulation scheme of Ω2(t)).

Further, when the modulation scheme applied to Ω1(t) is QPSK, the powerchanger (8501A) multiples Ω1(t) with a and thereby outputs a×Ω1(t). Onthe other hand, when the modulation scheme applied to Ω1(t) is 16QAM,the power changer (8501A) multiples Ω1(t) with b and thereby outputsb×Ω1(t).

Further, when the modulation scheme applied to Ω2(t) is QPSK, the powerchanger (8501B) multiples Ω2(t) with a and thereby outputs a×Ω2(t). Onthe other hand, when the modulation scheme applied to Ω2(t) is 16QAM,the power changer (8501B) multiples Ω2(t) with b and thereby outputsb×Ω2(t).

Note that, regarding the scheme for differently setting the averagepower of signals in accordance with mapping for QPSK modulation and theaverage power of signals in accordance with mapping for 16QAMmodulation, description has already been made in Embodiment F1.

Thus, when taking the set of (modulation scheme of Ω1(t), modulationscheme of Ω2(t)) into consideration, the period (cycle) for the phasechange and the switching between modulation schemes is 6=3×2 (where 3:the number of phase changing values prepared as phase changing valuesused in the scheme for regularly performing phase change afterprecoding, and 2: both (QPSK, 16QAM) and (16QAM, QPSK) can be the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)) for each of thephase changing values), as shown in FIG. 91.

As description has been made in the above, by making an arrangement suchthat both (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)), and such thatboth (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of (modulationscheme of Ω1(t), modulation scheme of Ω2(t)) with respect to each of thephase changing values prepared as phase changing values used in thescheme for regularly performing phase change, the following advantageouseffects are to be yielded. That is, even when differently setting theaverage power of signals in accordance with mapping for QPSK modulationand the average power of signals in accordance with mapping for 16QAMmodulation, the influence with respect to the PAPR of the transmissionpower amplifier included in the transmission device is suppressed to aminimal extent, and thus the influence with respect to the powerconsumption of the transmission device is suppressed to a minimalextent, while the reception quality of data received by the receptiondevice in the LOS environment is improved, as explanation has alreadybeen provided in the present description.

Note that, although description has been provided in the above, takingas an example a case where the set of (modulation scheme of Ω1(t),modulation scheme of Ω2(t)) is (QPSK, 16QAM) and (16QAM, QPSK), thepossible sets of (modulation scheme of Ω1(t), modulation scheme ofΩ2(t)) are not limited to this. More specifically, the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)) may be one of:(QPSK, 64QAM), (64QAM, QPSK); (16QAM, 64QAM), (64QAM, 16QAM); (128QAM,64QAM), (64QAM, 128QAM); (256QAM, 64QAM), (64QAM, 256QAM), and the like.That is, the present invention is to be similarly implemented providedthat two different modulation schemes are prepared, and a different oneof the modulation schemes is applied to each of Ω1(t) and Ω2(t).

In FIG. 92, a chart is provided indicating the modulation scheme, thepower changing value, and the phase changing value to be set at each oftimes t=0 through t=11 with respect to the structural exampleillustrated in FIG. 90. Note that the chart in FIG. 92 differs from thechart in FIG. 91. Note that, concerning the modulated signals z1(t) andz2(t), the modulated signals z1(t) and z2(t) at the same time point areto be simultaneously transmitted from different transmit antennas at thesame frequency. (Although the chart in FIG. 92 is based on the timedomain, when using a multi-carrier transmission scheme as the OFDMscheme, switching between schemes may be performed according to thefrequency (subcarrier) domain, rather than according to the time domain.In such a case, replacement should be made of t=0 with f=f0, t=1 withf=f1, . . . , as is shown in FIG. 92. (Note that here, f denotesfrequencies (subcarriers), and thus, f0, f1, . . . , indicate differentfrequencies (subcarriers) to be used.) Further, note that concerning themodulated signals z1(f) and z2(f) in such a case, the modulated signalsz1(f) and z2(f) having the same frequency are to be simultaneouslytransmitted from different transmit antennas.

As illustrated in FIG. 92, when the modulation scheme applied is QPSK,the power changer (although referred to as the power changer herein, mayalso be referred to as an amplification changer or a weight unit)multiplies a (a being a real number) with respect to a signal modulatedin accordance with QPSK. Similarly, when the modulation scheme appliedis 16QAM, the power changer (although referred to as the power changerherein, may also be referred to as the amplification changer or theweight unit) multiplies b (b being a real number) with respect to asignal modulated in accordance with 16QAM.

In the example illustrated in FIG. 92, three phase changing values,namely y[0], y[1], and y[2] are prepared as phase changing values usedin the scheme for regularly performing phase change after precoding.Additionally, the period (cycle) for the scheme for regularly performingphase change after precoding is 3 (thus, each of t0-t2, t3-t5, . . .composes one period (cycle)).

Further, the modulation scheme applied to s1(t) is fixed to QPSK, andthe modulation scheme to be applied to s2(t) is fixed to 16QAM.Additionally, the signal switcher (9001) illustrated in FIG. 90 receivesthe mapped signals 307A and 307B and the control signal (8500) as inputthereto. The signal switcher (9001) performs switching with respect tothe mapped signals 307A and 307B according to the control signal (8500)(there are also cases where the switching is not performed), and outputsswitched signals (9002A: Ω1(t), and 9002B: Ω2(t)).

Here, it is important that:

When performing phase change according to y[0], both (QPSK, 16QAM) and(16QAM, QPSK) can be the set of (modulation scheme of Ω1(t), modulationscheme of Ω2(t)), when performing phase change according to y[1], both(QPSK, 16QAM) and (16QAM, QPSK) can be the set of (modulation scheme ofΩ1(t), modulation scheme of Ω2(t)), and similarly, when performing phasechange according to y[2], both (QPSK, 16QAM) and (16QAM, QPSK) can bethe set of (modulation scheme of Ω1(t), modulation scheme of Ω2(t)).

Further, when the modulation scheme applied to Ω1(t) is QPSK, the powerchanger (8501A) multiples Ω1(t) with a and thereby outputs a×Ω1(t). Onthe other hand, when the modulation scheme applied to Ω1(t) is 16QAM,the power changer (8501A) multiples 21(t) with b and thereby outputsb×Ω1(t).

Further, when the modulation scheme applied to Ω2(t) is QPSK, the powerchanger (8501B) multiples Ω2(t) with a and thereby outputs a×Ω2(t). Onthe other hand, when the modulation scheme applied to Ω2(t) is 16QAM,the power changer (8501B) multiples Ω2(t) with b and thereby outputsb×Ω2(t).

Note that, regarding the scheme for differently setting the averagepower of signals in accordance with mapping for QPSK modulation and theaverage power of signals in accordance with mapping for 16QAMmodulation, description has already been made in Embodiment F1.

Thus, when taking the set of (modulation scheme of Ω1(t), modulationscheme of Ω2(t)) into consideration, the period (cycle) for the phasechange and the switching between modulation schemes is 6=3×2 (where 3:the number of phase changing values prepared as phase changing valuesused in the scheme for regularly performing phase change afterprecoding, and 2: both (QPSK, 16QAM) and (16QAM, QPSK) can be the set of(modulation scheme of δ1(t), modulation scheme of Ω2(t)) for each of thephase changing values), as shown in FIG. 92.

As description has been made in the above, by making an arrangement suchthat both (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)), and such thatboth (QPSK, 16QAM) and (16QAM, QPSK) exist as the set of (modulationscheme of Ω1(t), modulation scheme of Ω2(t)) with respect to each of thephase changing values prepared as phase changing values used in thescheme for regularly performing phase change, the following advantageouseffects are to be yielded. That is, even when differently setting theaverage power of signals in accordance with mapping for QPSK modulationand the average power of signals in accordance with mapping for 16QAMmodulation, the influence with respect to the PAPR of the transmissionpower amplifier included in the transmission device is suppressed to aminimal extent, and thus the influence with respect to the powerconsumption of the transmission device is suppressed to a minimalextent, while the reception quality of data received by the receptiondevice in the LOS environment is improved, as explanation has alreadybeen provided in the present description.

Note that, although description has been provided in the above, takingas an example a case where the set of (modulation scheme of Ω1(t),modulation scheme of Ω2(t)) is (QPSK, 16QAM) and (16QAM, QPSK), thepossible sets of (modulation scheme of Ω1(t), modulation scheme ofΩ2(t)) are not limited to this. More specifically, the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)) may be one of:(QPSK, 64QAM), (64QAM, QPSK); (16QAM, 64QAM), (64QAM, 16QAM); (128QAM,64QAM), (64QAM, 128QAM); (256QAM, 64QAM), (64QAM, 256QAM), and the like.That is, the present invention is to be similarly implemented providedthat two different modulation schemes are prepared, and a different oneof the modulation schemes is applied to each of Ω1(t) and Ω2(t).

Additionally, the relation between the modulation scheme, the powerchanging value, and the phase changing value set at each of times (orfor each of frequencies) is not limited to those described in the abovewith reference to FIGS. 91 and 92.

To summarize the explanation provided in the above, the following pointsare essential.

Arrangements are to be made such that the set of (modulation scheme ofΩ1(t), modulation scheme of Ω2(t)) can be either (modulation scheme A,modulation scheme B) or (modulation scheme B, modulation scheme A), andsuch that the average power for the modulation scheme A and the averagepower for the modulation scheme B are differently set.

Further, when the modulation scheme applied to ω1(t) is modulationscheme A, the power changer (8501A) multiples Ω1(t) with a and therebyoutputs a×Ω1(t). On the other hand, when the modulation scheme appliedto Ω1(t) is modulation scheme B, the power changer (8501A) multiplesΩ1(t) with b and thereby outputs b×Ω1(t). Further, when the modulationscheme applied to Ω2(t) is modulation scheme A, the power changer(8501B) multiples Ω2(t) with a and thereby outputs a×Ω2(t). On the otherhand, when the modulation scheme applied to Ω2(t) is modulation schemeB, the power changer (8501B) multiples Ω2(t) with b and thereby outputsb×Ω2(t).

Further, an arrangement is to be made such that phase changing valuesy[0], y[1], . . . , y[n−2], and y[n−1] (or y[k], where k satisfies0≤k≤n−1) exist as phase changing values prepared for use in the schemefor regularly performing phase change after precoding. Further, anarrangement is to be made such that both (modulation scheme A,modulation scheme B) and (modulation scheme B, modulation scheme A)exist as the set of (modulation scheme of Ω1(t), modulation scheme ofΩ2(t)) for y[k]. (Here, the arrangement may be made such that both(modulation scheme A, modulation scheme B) and (modulation scheme B,modulation scheme A) exist as the set of (modulation scheme of Ω1(t),modulation scheme of Ω2(t)) for y[k] for all values of k, or such that avalue k exists where both (modulation scheme A, modulation scheme B) and(modulation scheme B, modulation scheme A) exist as the set of(modulation scheme of Ω1(t), modulation scheme of Ω2(t)) for y[k].)

As description has been made in the above, by making an arrangement suchthat both (modulation scheme A, modulation scheme B) and (modulationscheme B, modulation scheme A) exist as the set of (modulation scheme ofΩ1(t), modulation scheme of Ω2(t)), and such that both (modulationscheme A, modulation scheme B) and (modulation scheme B, modulationscheme A) exist as the set of (modulation scheme of Ω1(t), modulationscheme of Ω2(t)) with respect to each of the phase changing valuesprepared as phase changing values used in the scheme for regularlyperforming phase change, the following advantageous effects are to beyielded. That is, even when differently setting the average power ofsignals for modulation scheme A and the average power of signals formodulation scheme B, the influence with respect to the PAPR of thetransmission power amplifier included in the transmission device issuppressed to a minimal extent, and thus the influence with respect tothe power consumption of the transmission device is suppressed to aminimal extent, while the reception quality of data received by thereception device in the LOS environment is improved, as explanation hasalready been provided in the present description.

Subsequently, explanation is provided of the operations of the receptiondevice. Explanation of the reception device has already been provided inEmbodiment 1 and so on, and the structure of the reception device isillustrated in FIGS. 7, 8 and 9, for instance.

According to the relation illustrated in FIG. 5, when the transmissiondevice transmits modulated signals as introduced in FIGS. 87, 88, 91 and92, one relation among the two relations denoted by the two formulasbelow is satisfied. Note that in the two formulas below, r1(t) and r2(t)indicate reception signals, and h11(t), h12(t), h21(t), and h22(t)indicate channel fluctuation values.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 93} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{{r2}(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}a & 0 \\0 & b\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu}{G1}} \right) \\\left\lbrack {{Math}.\mspace{14mu} 94} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{r\; 1(t)} \\{{r2}(t)}\end{pmatrix} = {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}}} \\{\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}{h\; 11(t)} & {h\; 12(t)} \\{h\; 21(t)} & {h\; 22(t)}\end{pmatrix}\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}b & 0 \\0 & a\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu}{G2}} \right)\end{matrix}$

Note that F shown in formulas G1 and G2 denotes precoding matrices usedat time t, and y(t) denotes phase changing values. The reception deviceperforms demodulation (detection) of signals by utilizing the relationdefined in the two formulas above (that is, demodulation is to beperformed in the same manner as explanation has been provided inEmbodiment 1). However, the two formulas above do not take intoconsideration such distortion components as noise components, frequencyoffsets, and channel estimation errors, and thus, the demodulation(detection) is performed with such distortion components included in thesignals. Regarding the values u and v that the transmission device usesfor performing the power change, the transmission device transmitsinformation about these values, or transmits information of thetransmission mode (such as the transmission scheme, the modulationscheme and the error correction scheme) to be used. The reception devicedetects the values used by the transmission device by acquiring theinformation, obtains the two formulas described above, and performs thedemodulation (detection).

Although description is provided in the present invention taking as anexample a case where switching between phase changing values isperformed in the time domain, the present invention may be similarlyembodied when using a multi-carrier transmission scheme such as OFDM orthe like and when switching between phase changing values in thefrequency domain, as description has been made in other embodiments. Ifthis is the case, t used in the present embodiment is to be replacedwith f (frequency ((sub) carrier)). Further, the present invention maybe similarly embodied in a case where switching between phase changingvalues is performed in the time-frequency domain. In addition, in thepresent embodiment, the scheme for regularly performing phase changeafter precoding is not limited to the scheme for regularly performingphase change after precoding, explanation of which has been provided inthe other sections of the present description. Further in addition, thesame effect of minimalizing the influence with respect to the PAPR is tobe obtained when applying the present embodiment with respect to aprecoding scheme where phase change is not performed.

Embodiment G2

In the present embodiment, description is provided on the scheme forregularly performing phase change, the application of which realizes anadvantageous effect of reducing circuit size when the broadcast (orcommunications) system supports both of a case where the modulationscheme applied to s1 is QPSK and the modulation scheme applied to s2 is16QAM, and a case where the modulation scheme applied to s1 is 16QAM andthe modulation scheme applied to s2 is 16QAM.

Firstly, explanation is made of the scheme for regularly performingphase change in a case where the modulation scheme applied to s1 is16QAM and the modulation scheme applied to s2 is 16QAM.

Examples of the precoding matrices used in the scheme for regularlyperforming phase change in a case where the modulation scheme applied tos1 is 16QAM and the modulation scheme applied to s2 is 16QAM are shownin Embodiment 1. The precoding matrices [F] are represented as follows.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 95} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\; 0}} \\{\alpha \times e^{j\; 0}} & e^{j\;\pi}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G3}} \right)\end{matrix}$

In the following, description is provided on an example where theformula G3 is used as the precoding matrices for the scheme forregularly performing phrase change after precoding in a case where 16QAMis applied as the modulation scheme to both s1 and s2.

FIG. 93 illustrates a structural example of the periphery of theweighting unit (precoding unit) which supports both of a case where themodulation scheme applied to s1 is QPSK and the modulation schemeapplied to s2 is 16QAM, and a case where the modulation scheme appliedto s1 is 16QAM and the modulation scheme applied to s2 is 16QAM. In FIG.93, elements that operate in a similar way to FIG. 3, FIG. 6 and FIG. 85bear the same reference signs, and explanations thereof are omitted.

In FIG. 93, the baseband signal switcher 9301 receives the precodedsignal 309A(z1(t)), the precoded and phase-changed signal 309B(z2(t)),and the control signal 8500 as input. When the control signal 8500indicates “do not perform switching of signals”, the precoded signal309A(z1(t)) is output as the signal 9302A(z1′(t)), and the precoded andphase-changed signal 309B(z2(t)) is output as the signal 9302B(z2′(t)).

In contrast, when the control signal 8500 indicates “perform switchingof signals”, the baseband signal switcher 8501 performs the following:

When time is 2k (k is an integer),

outputs the precoded signal 309A(z1(2k)) as the signal 9302A(r1(2k)),and outputs the precoded signal 309B(z2(2k)) as the precoded andphase-changed signal 9302B(r2(2k)),

When time is 2k+1 (k is an integer),

outputs the precoded and phase-changed signal 309B(z2(2k+1)) as thesignal 9302A(r1(2k+1)), and outputs the precoded signal 309A(z1(2k+1))as the signal 9302B(r2(2k+1)), and further,

When time is 2k (k is an integer),

outputs the precoded signal 309B(z2(2k)) as the signal 9302A(r1(2k)),and outputs the precoded signal 309A(z1(2k)) as the precoded andphase-changed signal 9302B(r2(2k)),

When time is 2k+1 (k is an integer),

outputs the precoded signal 309A(z1(2k+1)) as the signal9302A(r1(2k+1)), and outputs the precoded and phase-changed signal309B(z2(2k+1)) as the signal 9302B(r2(2k+1)). (Although the abovedescription provides an example of the switching between signals, theswitching between signals is not limited to this. It is to be noted thatimportance lies in that switching between signals is performed when thecontrol signal indicates “perform switching of signals”.)

As explained in FIG. 3, FIG. 4, FIG. 5, FIG. 12, FIG. 13 and so on, thesignal 9302A(r1(t)) is transmitted from an antenna instead of z1(t)(Note that predetermined processing is performed as shown in FIG. 3,FIG. 4, FIG. 5, FIG. 12, FIG. 13 and so on). Also, the signal9302B(r2(t)) is transmitted from an antenna instead of z2(t) (Note thatpredetermined processing is performed as shown in FIG. 3, FIG. 4, FIG.5, FIG. 12, FIG. 13 and so on.) Note that the signal 9302A(r1(t)) andthe signal 9302B(r2(t)) are transmitted from different antenna.

Here, it should be noted that the switching of signals as described inthe above is performed with respect to only precoded symbols. That is,the switching of signals is not performed with respect to other insertedsymbols such as pilot symbols and symbols for transmitting informationthat is not to be procoded (e.g. control information symbols), forexample. Further, although the description is provided in the above of acase where the scheme for regularly performing phase change afterprecoding is applied in the time domain, the present invention is notlimited to this. The present embodiment may be similarly applied also incases where the scheme for regularly performing phase change afterprecoding is applied in the frequency domain and in the time-frequencydomain. Similarly, the switching of signals may be performed in thefrequency domain or the time-frequency domain, even though descriptionis provided in the above where switching of signals is performed in thetime domain.

Subsequently, explanation is provided concerning the operation of eachof the units in FIG. 93 in a case where 16QAM is applied as themodulation scheme for both s1 and s2.

Since s1(t) and s2(t) are baseband signals (mapped signals) mapped withthe modulation scheme 16QAM, the mapping scheme applied thereto is asillustrated in FIG. 80, and g is represented by formula 79.

The power changer (8501A) receives a (mapped) baseband signal 307A forthe modulation scheme 16QAM and the control signal (8500) as input.Letting a value for power change set based on the control signal (8500)be v, the power changer outputs a signal (power-changed signal: 8502A)obtained by multiplying the (mapped) baseband signal 307A for themodulation scheme 16QAM by v.

The power changer (8501B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8500) as input.Letting a value for power change set based on the control signal (8500)be u, the power changer outputs a signal (power-changed signal: 8502B)obtained by multiplying the (mapped) baseband signal 307B for themodulation scheme 16QAM by u.

Here, the factors v and u satisfy: v=u=Ω, v²:u²=1:1. By making such anarrangement, data is received at an excellent reception quality by thereception device.

The weighting unit 600 receives the power-changed signal 8502A (thesignal obtained by multiplying the baseband signal (mapped signal) 307Amapped with the modulation scheme 16QAM by the factor v), thepower-changed signal 8502B (the signal obtained by multiplying thebaseband signal (mapped signal) 307B mapped with the modulation scheme16QAM by the factor u) and the information 315 regarding the weightingscheme as input. Further, the weighting unit 600 determines theprecoding matrix based on the information 315 regarding the weightingscheme, and outputs the precoded signal 309A(z1(t)) and the precodedsignal 316B(z2′(t)).

The phase changer 317B performs phase change on the precoded signal316B(z2′(t)), based on the information 315 regarding the informationprocessing scheme, and outputs the precoded and phase-changed signal309B(z2(t)).

Here, when F represents a precoding matrix used in the scheme forregularly performing phase change after precoding and y(t) representsthe phase changing values, the following formula holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 96} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}\Omega & 0 \\0 & \Omega\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu}{G4}} \right)\end{matrix}$

Note that y(t) is an imaginary number having the absolute value of 1(i.e. y[i]=ejθ).

When the precoding matrix F, which is a precoding matrix used in thescheme for regularly performing phase change after precoding, isrepresented by formula G3 and when 16QAM is applied as the modulationscheme of both s1 and s2, formula 37 is suitable as the value of α, asis described in Embodiment 1. When α is represented by formula 37, z1(t)and z2(t) each are baseband signals corresponding to one of the 256signal points in the IQ plane, as illustrated in FIG. 94. Note that FIG.94 illustrates an example of the layout of the 256 signal points, andthe layout may be a phase-rotated layout of the 256 signal components.

Here, since the modulation scheme applied to s1 is 16QAM and themodulation scheme applied to s2 is also 16QAM, the weighted andphase-changed signals z1(t) and z2(t) are each transmitted as 4 bitsaccording to 16QAM. Therefore a total of 8 bits are transferred as isindicated by the 256 signals points illustrated in FIG. 94. In such acase, since the minimum Euclidian distance between the signal points iscomparatively large, the reception quality of data received by thereception unit is improved.

The baseband signal switcher 9301 receives the precoded signal309A(z1(t)), the precoded and phase-changed signal 309B(z2(t)), and thecontrol signal 8500 as input. Since 16QAM is applied as the modulationscheme of both s1 and s2, the control signal 8500 indicates “do notperform switching of signals”. Thus, the precoded signal 309A(z1(t)) isoutput as the signal 9302A(r1(t)) and the precoded and phase-changedsignal 309B(z2(t)) is output as the signal 9302B(r2(t)).

Subsequently, explanation is provided concerning the operation of eachof the units in FIG. 116 in a case where QPSK is applied as themodulation scheme for s1 and 16QAM is applied as the modulation schemefor s2.

Let s1(t) be the (mapped) baseband signal for the modulation schemeQPSK. The mapping scheme for s1(t) is as shown in FIG. 81, and h is asrepresented by formula 78. Since s2(t) is the (mapped) baseband signalfor the modulation scheme 16QAM, the mapping scheme for s2(t) is asshown in FIG. 80, and g is as represented by formula 79.

The power changer (8501A) receives the baseband signal (mapped signal)307A mapped according to the modulation scheme QPSK, and the controlsignal (8500) as input. Further, the power changer (8501A) multipliesthe baseband signal (mapped signal) 307A mapped according to themodulation scheme QPSK by a factor v, and outputs the signal obtained asa result of the multiplication (the power-changed signal: 8502A). Thefactor v is a value for performing power change and is set according tothe control signal (8500).

The power changer (8501B) receives a (mapped) baseband signal 307B forthe modulation scheme 16QAM and a control signal (8500) as input.Letting a value for power change set based on the control signal (8500)be u, the power changer outputs a signal (power-changed signal: 8502B)obtained by multiplying the (mapped) baseband signal 307B for themodulation scheme 16QAM by u.

In Embodiment F1, description is provided that one exemplary example iswhere “the ratio between the average power of QPSK and the average powerof 16QAM is set so as to satisfy the formula v²:u²=1:5”. (By making suchan arrangement, data is received at an excellent reception quality bythe reception device.) In the following, explanation is provided of thescheme for regularly performing phase change after precoding when suchan arrangement is made.

The weighting unit 600 receives the power-changed signal 8502A (thesignal obtained by multiplying the baseband signal (mapped signal) 307Amapped with the modulation scheme QPSK by the factor v), thepower-changed signal 8502B (the signal obtained by multiplying thebaseband signal (mapped signal) 307B mapped with the modulation scheme16QAM by the factor u) and the information 315 regarding the signalprocessing scheme as input. Further, the weighting unit 600 performsprecoding according to the the information 315 regarding the signalprocessing scheme, and outputs the precoded signal 309A(z1(t)) and theprecoded signal 316B(z2′(t)).

Here, when F represents a precoding matrix used in the scheme forregularly performing phase change after precoding and y(t) representsthe phase change values, the following formula holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 97} \right\rbrack & \; \\\begin{matrix}{\begin{pmatrix}{z\; 1(t)} \\{z\; 2(t)}\end{pmatrix} = {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}{ve}^{j\; 0} & 0 \\0 & {ue}^{j\; 0}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & u\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}} \\{= {\begin{pmatrix}1 & 0 \\0 & {y(t)}\end{pmatrix}{F\begin{pmatrix}v & 0 \\0 & {\sqrt{5}v}\end{pmatrix}}\begin{pmatrix}{s\; 1(t)} \\{s\; 2(t)}\end{pmatrix}}}\end{matrix} & \left( {{formula}\mspace{14mu}{G5}} \right)\end{matrix}$

Note that y(t) is an imaginary number having the absolute value of 1(i.e. y[i]=e^(jθ)).

When the precoding matrix F, which is a precoding matrix according tothe precoding scheme for regularly performing phase change afterprecoding, is represented by formula G3 and when 16QAM is applied as themodulation scheme of both s1 and s2, formula 37 is suitable as the valueof a, as is described. The reason for this is explained in thefollowing.

FIG. 95 illustrates the relationship between the 16 signal points of16QAM and the 4 signal points of QPSK on the IQ plane when thetransmission state is as described in the above. In FIG. 95, each ∘indicates a signal point of 16QAM, and each ● indicates a signal pointof QPSK. As can be seen in FIG. 95, four signal points among the 16signal points of the 16QAM coincide with the 4 signal points of theQPSK. Under such circumstances, when the precoding matrix F, which is aprecoding matrix used in the scheme for regularly performing phasechange after precoding, is represented by formula G3 and when formula 37is the value of a, each of z1(t) and z2(t) is a baseband signalcorresponding to 64 signal points extracted from the 256 signal pointsillustrated in FIG. 94 of a case where the modulation scheme applied tos1 is 16QAM and the modulation scheme applied to s2 is 16QAM. Note thatFIG. 94 illustrates an example of the layout of the 256 signal points,and the layout may be a phase-rotated layout of the 256 signalcomponents.

Since QPSK is the modulation scheme applied to s1 and 16QAM is themodulation scheme applied to s2, the weighted and phase-changed signalsz1(t) and z2(t) are respectively transmitted as 2 bits according toQPSK, and 4 bits according to 16QAM. Therefore a total of 6 bits aretransferred as is indicated by the 64 signals points. Since the minimumEuclidian distance between the 64 signal points as described in theabove is comparatively large, the reception quality of the data receivedby the reception device is improved.

The baseband signal switcher 9301 receives the precoded signal309A(z1(t)), the precoded and phase-changed signal 309B(z2(t)), and thecontrol signal 8500 as input. Since QPSK is the modulation scheme for s1and 16QAM is the modulation scheme for s2 and thus, the control signal8500 indicates “perform switching of signals”, the baseband signalswitcher 9301 performs, for instance, the following:

When time is 2k (k is an integer),

outputs the precoded signal 309A(z1(2k)) as the signal 9302A(r1(2k)),and outputs the precoded signal 309B(z2(2k)) as the precoded andphase-changed signal 9302B(r2(2k)),

When time is 2k+1 (k is an integer),

outputs the precoded and phase-changed signal 309B(z2(2k+1)) as thesignal 9302A(r1(2k+1)), and outputs the precoded signal 309A(z1(2k+1))as the signal 9302B(r2(2k+1)), and further,

When time is 2k (k is an integer),

outputs the precoded signal 309B(z2(2k)) as the signal 9302A(r1(2k)),and outputs the precoded signal 309A(z1(2k)) as the precoded andphase-changed signal 9302B(r2(2k)),

When time is 2k+1 (k is an integer),

outputs the precoded signal 309A(z1(2k+1)) as the signal9302A(r1(2k+1)), and outputs the precoded and phase-changed signal309B(z2(2k+1)) as the signal 9302B(r2(2k+1)).

Note that, in the above, description is made that switching of signalsis performed when QPSK is the modulation scheme applied to s1 and 16QAMis the modulation scheme applied to s2. By making such an arrangement,the reduction of PAPR is realized and further, the electric consumptionby the transmission unit is suppressed, as description has been providedin Embodiment F1. However, when the electric consumption by thetransmission device need not be taken into account, an arrangement maybe made such that switching of signals is not performed similar to thecase where 16QAM is applied as the modulation scheme for both s1 and s2.

Additionally, description has been provided in the above on a case whereQPSK is the modulation scheme applied to s1 and 16QAM is the modulationscheme applied to s2, and further, the condition v²:u²=1:5 is satisfied,since such a case is considered to be exemplary. However, there exists acase where excellent reception quality is realized when (i) the schemefor regularly performing phase change after precoding when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2 and (ii) the scheme for regularly performing phase changeafter precoding when 16QAM is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 are considered as beingidentical under the condition v²<u². Thus, the condition to be satisfiedby values v and u is not limited to v²:u²=1:5.

By considering (i) the scheme for regularly performing phase changeafter precoding when QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 and (ii) the scheme forregularly performing phase change after precoding when 16QAM is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2 to be identical as explained in the above, the reductionof circuit size is realized. Further, in such a case, the receptiondevice performs demodulation according to formulas G4 and G5, and to thescheme of switching between signals, and since signal points coincide asexplained in the above, the sharing of a single arithmetic unitcomputing reception candidate signal points is possible, and thus, thecircuit size of the reception device can be realized to a furtherextent.

Note that, although description has been provided in the presentembodiment taking the formula G3 as an example of the scheme forregularly performing phase change after precoding, the scheme forregularly performing phase change after precoding is not limited tothis.

The essential points of the present invention are as described in thefollowing:

When both the case where QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 and the case where 16QAM isthe modulation scheme applied for both s1 and s2 are supported, the samescheme for regularly performing phase change after precoding is appliedin both cases.

The condition v²=u² holds when 16QAM is the modulation scheme appliedfor both s1 and s2, and the condition v²<u² holds when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2

Further, examples where excellent reception quality of the receptiondevice is realized are described in the following.

Example 1 (the Following Two Conditions are to be Satisfied)

The condition v²=u² holds when 16QAM is the modulation scheme appliedfor both s1 and s2, and the condition v²:u²=1:5 holds when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2, and

The same scheme for regularly performing phase change after precoding isapplied in both of cases where 16QAM is the modulation scheme appliedfor both s1 and s2 and QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2.

Example 2 (the Following Two Conditions are to be Satisfied)

The condition v²=u² holds when 16QAM is the modulation scheme appliedfor both s1 and s2, and the condition v²<u² holds when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2, and

When both the case where QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 and the case where 16QAM isthe modulation scheme applied for both s1 and s2 are supported, the samescheme for regularly performing phase change after the precoding isapplied in both cases, and the precoding matrices are represented byformula G3.

Example 3 (the Following Two Conditions are to be Satisfied)

The condition v²=u² holds when 16QAM is the modulation scheme appliedfor both s1 and s2, and the condition v²<u² holds when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2, and

When both the case where QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 and the case where 16QAM isthe modulation scheme applied for both s1 and s2 are supported, the samescheme for regularly performing phase change after the precoding isapplied in both cases, and the precoding matrices are represented byformula G3, and α is represented by formula 37.

Example 4 (the Following Two Conditions are to be Satisfied)

The condition v²=u² holds when 16QAM is the modulation scheme appliedfor both s1 and s2, and the condition v²:u²=1:5 holds when QPSK is themodulation scheme applied to s1 and 16QAM is the modulation schemeapplied to s2.

When both the case where QPSK is the modulation scheme applied to s1 and16QAM is the modulation scheme applied to s2 and the case where 16QAM isthe modulation scheme applied for both s1 and s2 are supported, the samescheme for regularly performing phase change after the precoding isapplied in both cases, and the precoding matrices are represented byformula G3, and a is represented by formula 37.

Note that, although the present embodiment has been described with anexample where the modulation schemes are QPSK and 16QAM, the presentembodiment is not limited to this example. The scope of the presentembodiment may be expanded as described below. Consider a modulationscheme A and a modulation scheme B. Let a be the number of a signalpoint on the IQ plane of the modulation scheme A, and let b be thenumber of signal points on the IQ plane of the modulation scheme B,where a<b. Then, the essential points of the present invention aredescribed as follows.

The following two conditions are to be satisfied.

If the case where the modulation scheme of s1 is the modulation scheme Aand the modulation scheme of s2 is the modulation scheme B, and the casewhere the modulation scheme of s1 is the modulation scheme B and themodulation scheme of s2 is the modulation scheme B are both supported,the same scheme is used in common in both the cases for for regularlyperforming phase change after precoding.

When the modulation scheme of s1 is the modulation scheme B and themodulation scheme of s2 is the modulation scheme B, the condition v²=u²is satisfied, and when the modulation scheme of s1 is the modulationscheme A and the modulation scheme of s2 is the modulation scheme B, thecondition v²<u² is satisfied.

Here, the baseband signal switching as described with reference to FIG.93 may be optionally executed. However, when the modulation scheme of s1is the modulation scheme A and the modulation scheme of s2 is themodulation scheme B, it is preferable to perform the above-describedbaseband signal switching with the influence of the PAPR taken intoaccount.

Alternatively, the following two conditions are to be satisfied.

If the case where the modulation scheme of s1 is the modulation scheme Aand the modulation scheme of s2 is the modulation scheme B, and the casewhere the modulation scheme of s1 is the modulation scheme B and themodulation scheme of s2 is the modulation scheme B are both supported,the same scheme is used in common in both the cases for for regularlyperforming phase change after precoding, and the precoding matrices arepresented by formula G3.

When the modulation scheme of s1 is the modulation scheme B and themodulation scheme of s2 is the modulation scheme B, the condition v²=u²is satisfied, and when the modulation scheme of s1 is the modulationscheme A and the modulation scheme of s2 is the modulation scheme B, thecondition v²<u² is satisfied.

Here, the baseband signal switching as described with reference to FIG.93 may be optionally executed. However, when the modulation scheme of s1is the modulation scheme A and the modulation scheme of s2 is themodulation scheme B, it is preferable to perform the above-describedbaseband signal switching with the influence of the PAPR taken intoaccount.

As an exemplary set of the modulation scheme A and the modulation schemeB, (modulation scheme A, modulation scheme B) is one of (QPSK, 16QAM),(16QAM, 64QAM), (64QAM, 128QAM), and (64QAM, 256QAM).

Although the above explanation is given for an example where phasechange is performed on one of the signals after precoding, the presentinvention is not limited to this. As described in this Description, evenwhen phase change is performed on a plurality of precoded signals, thepresent embodiment is applicable. If this is the case, the relationshipbetween the modulated signal set and the precoding matrices (theessential points of the present invention).

Further, although the present embodiment has been described on theassumption that the precoding matrices F are represented by formula G3,the present invention is not limited to this. For example, any one ofthe following may be used:

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 98} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}{\alpha \times e^{j\; 0}} & e^{j\;\pi} \\e^{j\; 0} & {\alpha \times e^{j\; 0}}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G6}} \right) \\\left\lbrack {{Math}\;.\mspace{11mu} 99} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\; 0} & {\alpha \times e^{j\;\pi}} \\{\alpha \times e^{j\; 0}} & e^{j\; 0}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G7}} \right) \\\left\lbrack {{Math}.\mspace{14mu} 100} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}{\alpha \times e^{j\; 0}} & e^{j\; 0} \\e^{j\; 0} & {\alpha \times e^{j\;\pi}}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G8}} \right) \\\left\lbrack {{Math}.\mspace{14mu} 101} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}e^{j\;\theta_{11}} & {\alpha \times e^{j{({\theta_{11} + \lambda})}}} \\{\alpha \times e^{j\;\theta_{21}}} & e^{j{({\theta_{21} + \lambda + \pi})}}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G9}} \right) \\\left\lbrack {{Math}.\mspace{14mu} 102} \right\rbrack & \; \\{F = {\frac{1}{\sqrt{\alpha^{2} + 1}}\begin{pmatrix}{\alpha \times e^{j\;\theta_{11}}} & e^{j{({\theta_{11} + \lambda + \pi})}} \\e^{j\;\theta_{21}} & {\alpha \times e^{j{({\theta_{21} + \lambda})}}}\end{pmatrix}}} & \left( {{formula}\mspace{14mu}{G10}} \right)\end{matrix}$

Note that θ₁₁, θ₂₁ and λ in formulas G9 and G10 are fixed values(radians).

Although description is provided in the present invention taking as anexample a case where switching between phase change values is performedin the time domain, the present invention may be similarly embodied whenusing a multi-carrier transmission scheme such as OFDM or the like andwhen switching between phase change values in the frequency domain, asdescription has been made in other embodiments. If this is the case, tused in the present embodiment is to be replaced with f (frequency((sub) carrier)). Further, the present invention may be similarlyembodied in a case where switching between phase change values isperformed in the time-frequency domain. Note that, in the presentembodiment, the scheme for regularly performing phase change afterprecoding is not limited to the scheme for regularly performing phasechange after precoding as described in this Description.

Furthermore, in any one of the two patterns of setting the modulationscheme according to the present embodiment, the reception deviceperforms demodulation and detection using the reception scheme describedin Embodiment F1.

INDUSTRIAL APPLICABILITY

The present invention is widely applicable to wireless systems thattransmit different modulated signals from a plurality of antennas, suchas an OFDM-MIMO system. Furthermore, in a wired communication systemwith a plurality of transmission locations (such as a Power LineCommunication (PLC) system, optical communication system, or DigitalSubscriber Line (DSL) system), the present invention may be adapted toMIMO, in which case a plurality of transmission locations are used totransmit a plurality of modulated signals as described by the presentinvention. A modulated signal may also be transmitted from a pluralityof transmission locations.

REFERENCE SIGNS LIST

-   -   302A, 302B Encoders    -   304A, 304B Interleavers    -   306A, 306B Mappers    -   314 Signal processing scheme information generator    -   308A, 308B Weighting units    -   310A, 310B Wireless units    -   312A, 312B Antennas    -   317A, 317B Phase changers    -   402 Encoder    -   404 Distributor    -   504#1, 504#2 Transmit antennas    -   505#1, 505#2 Receive antennas    -   600 Weighting unit    -   701_X, 701_Y Antennas    -   703_X, 703_Y Wireless units    -   705_1 Channel fluctuation estimator    -   705_2 Channel fluctuation estimator    -   707_1 Channel fluctuation estimator    -   707_2 Channel fluctuation estimator    -   709 Control information decoder    -   711 Signal processor    -   803 Inner MIMO detector    -   805A, 805B Log-likelihood calculators    -   807A, 807B Deinterleavers    -   809A, 809B Log-likelihood ratio calculators    -   811A, 811B Soft-in/soft-out decoders    -   813A, 813B Interleavers    -   815 Memory    -   819 Coefficient generator    -   901 Soft-in/soft-out decoder    -   903 Distributor    -   1201A, 1201B OFDM-related processors    -   1302A, 1302A Serial-to-parallel converters    -   1304A, 1304B Reorderers    -   1306A, 1306B IFFT units    -   1308A, 1308B Wireless units

The invention claimed is:
 1. A signal generation method comprising:phase-changing baseband signals with respective phase changing patternsto generate respective phase-changed signals, each of the phase changingpatterns being different from each other; andinverse-fast-Fourier-transforming the phase-changed signals torespective orthogonal frequency division multiplexing (OFDM)transmission signals, wherein each phase changing pattern has Ncandidates for an amount of change in a phase, N being an integergreater than two, and each candidate is periodically selected from the Ncandidates based on subcarriers of the respective OFDM transmissionsignals, a phase of a corresponding baseband signal among the basebandsignals being changed by the each candidate.
 2. A signal generationapparatus comprising: phase change circuit configured to phase-changebaseband signals with respective phase changing patterns to generaterespective phase-changed signals, each of the phase changing patternsbeing different from each other; and inverse fast Fourier transformcircuit configured to inverse-fast-Fourier-transform the phase-changedsignals to respective orthogonal frequency division multiplexing (OFDM)transmission signals, wherein each phase changing pattern has Ncandidates for an amount of change in a phase, N being an integergreater than two, and each candidate is periodically selected from the Ncandidates based on subcarriers of the respective OFDM transmissionsignals, a phase of a corresponding baseband signal among the basebandsignals being changed by the each candidate.